Digital multilevel memory system having multistage autozero sensing

ABSTRACT

A digital multibit non-volatile memory integrated system includes autozero multistage sensing. One stage may provide local sensing with autozero. Another stage may provide global sensing with autozero. A twisted bitline may be used for array arrangement. Segment reference may be used for each segment. The system may read data cells using a current sensing one or two step binary search. The system may use inverse voltage mode or inverse current mode sensing. The system may use no current multilevel sensing. The system may use memory cell replica sensing. The system may use dynamic sensing. The system may use built-in byte redundancy. Sense amplifiers capable of sub-volt (&lt;&lt;1V) sensing are described.

CROSS-REFERENCE TO RELATED APPLICATIONS

This is a continuation-in-part of application Ser. No. 10/211,886, filedAug. 1, 2002, now U.S. Pat. No. 6,865,099 which is acontinuation-in-part of application Ser. No. 09/929,542, filed Aug. 13,2001, now U.S. Pat. No. 6,751,118 which is a division of applicationSer. No. 09/231,928 filed Jan. 14, 1999, issued as U.S. Pat. No.6,282,145, the subject matter of each of these applications isincorporated herein by reference.

This application is related to U.S. patent application Ser. No.10/317,455, filed on Dec. 11, 2002 herewith, entitled “MultistageAutozero Sensing For A Multilevel Non-volatile Memory Integrated CircuitSystem”, inventor Hieu Van Tran, the disclosure of which is incorporatedherein by reference, U.S. patent application Ser. No. 10/317,375,filedon Dec. 11, 2002 herewith, entitled “Digital Multilevel Non-VolatileMemory System”, inventor Hieu Van Tran, the disclosure of which isincorporated herein by reference, and U.S. patent application Ser. No.10/317,433, filed on Dec. 11, 2002 herewith, entitled “Sub-Volt Sensingfor Digital Multilevel Flash Memory”, inventor Hieu Van Tran, thedisclosure of which is incorporated herein by reference.

FIELD OF THE INVENTION

This invention relates in general to semiconductor memories, and, inparticular, to the design and operation of multilevel nonvolatilesemiconductor memories.

BACKGROUND OF THE INVENTION

As the information technology progresses, the demand for high densitygiga bit and tera bit memory integrated circuits is insatiable inemerging applications such as data storage for photo quality digitalfilm in multi-mega pixel digital camera, CD quality audio storage inaudio silicon recorder, portable data storage for instrumentation andportable personal computers, and voice, data, and video storage forwireless and wired phones and other personal communicating assistants.

The nonvolatile memory technology such as ROM (Read Only Memory), EEPROM(Electrical Erasable Programmable Read Only Memory), or FLASH is often atechnology of choice for these application due to its nonvolatilenature, meaning it still retains the data even if the power supplied toit is removed. This is in contrast with the volatile memory technology,such as DRAM (Dynamic Random Access Memory), which loses data if thepower supplied to it is removed. This nonvolatile feature is very usefulin saving the power from portable supplies, such as batteries. Untilbattery technology advances drastically to ensure typical electronicsystems to function for a typical operating lifetime, e.g., 10 years,the nonvolatile technology will fill the needs for most portableapplications.

The FLASH technology, due to its smallest cell size, is the highestdensity nonvolatile memory system currently available. The advance ofthe memory density is made possible by rapidly advancing the processtechnology into the realm of nano meter scale and possibly into theatomic scale and electron scale into the next century. At the presentsub-micro meter scale, the other method that makes the superhigh-density memory system possible is through the exploitation of theanalog nature of a storage element.

The analog nature of a flash or nonvolatile storage element provides, bytheory, an enormous capability to store information. For example, if oneelectron could represent one bit of information then, for one typicalconventional digital memory cell, the amount of information is equal tothe number of electrons stored, or approximately a few hundredthousands. Advances in device physics exploring the quantum mechanicalnature of the electronic structure will multiply the analog informationmanifested in the quantum information of a single electron even further.

The storage information in a storage element is hereby defined as adiscrete number of storage levels for binary digital signal processingwith the number of storage levels equal to 2^(N) with N equal to thenumber of digital binary bits. The optimum practical number of discretelevels stored in a nonvolatile storage element depends on the innovativecircuit design method and apparatus, the intrinsic and extrinsicbehavior of the storage element, all within constraints of a definiteperformance target, such as product speed and operating lifetime, with acertain cost penalty.

At the current state of the art, all the multilevel systems are onlysuitable for medium density, i.e. less than a few tens of mega bits, andonly suitable for a small number of storage levels per cell, i.e., lessthan four levels or two digital bits.

As can be seen, memories having high storage capacity and fast operatingspeed are highly desirable.

The signal path from the data cells to a sense amplifier may havemismatch with the signal path from the reference memory cells to thesense amplifier. The mismatch generates a current ratio error and may becaused by mismatches of the threshold voltage, the width, length,mobility, and oxide thickness of the circuit elements, such astransistors, in the signal paths. The mismatch also may be caused bymismatch in signal paths due to parasitics, such as width and length ofinterconnects.

SUMMARY OF THE INVENTION

A data storage system comprises a plurality of memory arrays. Eachmemory array comprises a plurality of memory subarrays that each includea plurality of data memory cells and a plurality of reference memorycells, and a plurality of local sense amplifiers. Each local senseamplifier is coupled to a corresponding one of the plurality of memorysubarrays and reads the contents of data memory cells by comparing thecontents to currents or voltages from reference memory cells within thecorresponding memory subarray. The local sense amplifier equalizes anoutput of the local sense amplifier to a current or voltage of thecorresponding reference memory cell prior to sensing of the data memorycell.

The data storage system may further comprise a plurality of global senseamplifiers that are each coupled to a group of the plurality of localsense amplifiers. The global sense amplifiers may include an autozerofunction to equalize an output of the global sense amplifier to an inputof the global sense amplifier prior to sensing of the data memory cell.

The memory subarrays may be arranged in pages of memory cells thatinclude both data memory cells and reference memory cells. The memorysystem may include segment reference cells.

The data storage system may include sub-volt sensing amplifiers andbuilt-in byte redundancy.

foregoing, together with other aspects of this invention, will becomemore apparent when referring to the following specification, claims, andaccompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1A is a cross section of a source side injection flash memory cell.

FIG. 1B is a transistor symbol corresponding to the source sideinjection flash memory cell shown in FIG. 1A.

FIG. 1C is a block diagram of a nonvolatile multilevel memory system.

FIG. 1D is a block diagram of an electronic camera system utilizing anonvolatile multilevel memory system.

FIG. 1E is a block diagram of an electronic audio system utilizing anonvolatile multilevel memory system.

FIG. 2A is a block diagram of super high-density nonvolatile multilevelmemory integrated circuit system.

FIG. 2B is a block diagram of flash power management unit.

FIG. 2C shows voltage mode sensing.

FIG. 3A is a block diagram of super high-density nonvolatile multilevelarray architecture.

FIG. 3B is a page select circuit, which together with the segment selectdecoder selects one bitline at a time for each y-driver.

FIG. 3C is a block diagram of a multilevel sub-array block.

FIG. 4A is one embodiment of a nonvolatile multilevel array unit ofinhibit and select segmentation.

FIG. 4B shows an alternate embodiment of the inhibit and selectsegmentation scheme.

FIG. 4C shows another alternate embodiment of the inhibit and selectsegmentation scheme.

FIG. 4D shows another alternate embodiment of the inhibit and selectsegmentation scheme.

FIG. 4E shows another alternate embodiment of the inhibit and selectsegmentation scheme.

FIG. 4F shows another alternate embodiment of the inhibit and selectsegmentation scheme.

FIG. 5A is a cross section of inhibit and select segmentationinterconnection.

FIG. 5B is a cross section of another embodiment of inhibit and selectsegmentation interconnection.

FIG. 5C is a 2-step ramp rate control and fast-slow ramp rate control.

FIG. 6 shows a block diagram of multilevel decoding.

FIG. 7 shows one segment decoder that includes segmented power supplydecoder, segmented bitline select decoder, inhibit decoder, segmentedpredecoded common line decoder, and control gate and control linedecoder.

FIG. 8 shows a segmented power supply decoder.

FIG. 9A shows a segmented bitline decoder.

FIG. 9B shows a segmented inhibit decoder.

FIG. 9C shows a segmented predecoded common line decoder.

FIG. 10 shows a sub-block decoder for control gate and common linemultilevel decoder.

FIG. 11A shows a sub-block of the circuit in FIG. 10 for four controlgates and one common line multilevel decoder.

FIG. 11B shows another embodiment of sub-block for four control gatesand one common line multilevel decoder with winner-take-all Kelvinconnection.

FIG. 11C shows a circuit for one common line driver.

FIG. 12 shows a scheme of the feedthrough-to-driver andfeedthrough-to-memory multilevel precision decoding.

FIG. 13 shows a block diagram of a multilevel reference system.

FIG. 14 shows details of a block diagram of a multilevel referencesystem.

FIG. 15 shows a reference detection scheme.

FIG. 16 shows positional linear reference system.

FIG. 17 shows a positional geometric reference system.

FIG. 18 shows an embodiment of geometric compensation reference scheme.

FIG. 19A shows voltage levels for program verify, margin, read, andrestore for one embodiment of the current invention.

FIG. 19B shows voltage levels for program verify, margin, read, andrestore for an alternative embodiment of the current invention.

FIG. 20 shows an embodiment of flow diagram of the page programmingcycle.

FIG. 21 shows an embodiment of flow diagram after page programmingbegins.

FIG. 22A shows a continuation of flow diagram after page programmingbegins.

FIG. 22B shows an alternative embodiment of continuation of flow diagramafter page programming begins shown in FIG. 22A.

FIG. 22C shows an alternate embodiment of the flow diagram shown in FIG.22B.

FIG. 23 shows an embodiment of flow diagram of the page read cycle.

FIG. 24 shows a continuation of flow diagram of the page read cycle inFIG. 23.

FIG. 25 shows a continuation of flow diagram of the page read cycle inFIG. 24.

FIG. 26 shows details of an embodiment of a single y-driver YDRVS 110S.

FIG. 27 shows details of a latch block, a program/read control block,and program/program inhibit block included in the single y-driver YDRVS110S.

FIG. 28 is a block diagram illustrating a memory system for a multilevelmemory.

FIG. 29A is a block diagram illustrating an inverter mode sensingcircuit.

FIG. 29B is a block diagram illustrating a voltage mode sensing circuit.

FIG. 30 is a block diagram illustrating a wide range, high speed voltagemode sensing circuit.

FIG. 31 is a block diagram illustrating a wide range, high speed modesensing circuit having a local source follower stage and a global commonsource stage.

FIG. 32 is a block diagram illustrating a wide range, high speed modesensing circuit with a local PMOS source follower stage and a globalsource follower stage.

FIG. 33 is a block diagram illustrating a wide range, high speed modesensing circuit with a local NMOS source follower stage and a globalsource following stage.

FIG. 34 is a block diagram illustrating a global sense amplifier havingan auto zeroing function.

FIG. 35 is a block diagram illustrating an auto zero sense amplifier.

FIG. 36 is a block diagram illustrating a memory system for a multilevelmemory including local autozero sense amplifiers and global autozerosense amplifiers.

FIG. 36A is a block diagram illustrating a memory system for amultilevel memory including local autozero sense amplifiers.

FIG. 37 is a block diagram illustrating a memory system including singleended autozero sense amplifiers.

FIG. 38 is a block diagram illustrating a memory system includingdifferential autozero sense amplifiers.

FIG. 39 is a block diagram illustrating a memory system includingcrossed bitlines.

FIG. 40 is a block diagram illustrating a current sense amplifierincluding an autozero.

FIG. 41 is a block diagram including a current sense amplifier includingautozero and replica loading.

FIG. 42 is a block diagram illustrating a two-stage current senseamplifier including autozero.

FIG. 43 is a block diagram illustrating a two-stage current senseamplifier including autozero.

FIG. 44 is a block diagram illustrating a two-stage indirect currentsense amplifier having autozero.

FIG. 45 is a block diagram illustrating a two-stage indirect currentsense amplifier having autozero.

FIG. 46 is a block diagram illustrating a memory system including a lowvoltage sense amplifier.

FIG. 46A is a block diagram illustrating a memory system including a lowvoltage sense amplifier.

FIG. 47 is a block diagram illustrating a memory system including a lowvoltage sense amplifier according to another embodiment.

FIG. 47A is a block diagram illustrating a memory system including a lowvoltage sense amplifier according to another embodiment.

FIG. 47B is a block diagram illustrating a memory system including a lowvoltage sense amplifier according to another embodiment.

FIG. 48 is a block diagram illustrating a memory system including a lowvoltage sense amplifier according to another embodiment.

FIG. 48A is a block diagram illustrating a memory system including a lowvoltage sense amplifier according to another embodiment.

FIG. 48B is a block diagram illustrating a memory system including a lowvoltage sense amplifier according to another embodiment.

FIG. 49 is a schematic diagram illustrating a shared sense amplifiersegmented reference array.

FIG. 50 is a schematic diagram illustrating a memory cell replica senseamplifier.

FIG. 51 is a schematic diagram illustrating a differential current senseamplifier.

FIG. 52 is a schematic diagram illustrating a two-stage differentialcurrent sense amplifier.

FIG. 53 is a schematic diagram illustrating a current difference senseamplifier.

FIG. 54 is a schematic diagram illustrating a current difference senseamplifier.

FIG. 55 is a schematic diagram illustrating a dynamic sense amplifier.

FIG. 56 is a graph illustrating control signals and voltage levels ofthe dynamic sense amplifier of FIG. 55.

FIG. 57 is a schematic diagram illustrating the dynamic charge senseamplifier.

FIG. 58 is a flow diagram illustrating a single bit current sensingbinary search.

FIG. 59 is a flow diagram illustrating a multiple bit current sensingbit search.

FIG. 60 is a block diagram illustrating a memory system with a built-inconcurrent byte redundancy.

DESCRIPTION OF THE SPECIFIC EMBODIMENTS

Described are the design method and apparatus for a super high densitynonvolatile memory system capable of giga to tera bits as applied to thearray architecture, reference system, and decoding schemes to realizethe optimum possible number of storage levels within specifiedperformance constraints. Method and apparatus for multilevel program andsensing algorithm and system applied to flash memory is also described.

Array architectures and operating methods are described that aresuitable for a super high density, in the giga to tera bits, formultilevel nonvolatile “green” memory integrated circuit system. “Green”refers to a system working in an efficient and low power consumptionmanner. The system and method solves the issues associated with superhigh density multilevel memory system, such as, precision voltagecontrol in the array, severe capacitive loading from MOS transistorgates and parasitics, high leakage current due to memory cells and fromcells to cells, excessive power consumption due to large number of gatesand parasitics, and excessive memory cell disturbances due to largememory density.

An Inhibit and Select Segmentation Scheme uses a truly-floating-bitlinescheme to greatly reduce the capacitance from junctions and parasiticinterconnects to a small value.

A Multilevel Memory Decoding scheme is capable of greater than 10-bitmultilevel operation. The Multilevel Memory Decoding Scheme includes thePower Supply Decoded Decoding Scheme, the Feedthrough-to-Memory DecodingScheme, and the Feedthrough-to-Driver Decoding Scheme. The MultilevelMemory Decoding scheme also includes a “winner-take-all” Kelvin DecodingScheme, which provides precise bias levels for the memory at a minimumcost. A constant-total-current-program scheme is described. Fast-slowand 2-step ramp rate control programming are described. A referencesystem method and apparatus includes the Positional Linear ReferenceSystem, Positional Geometric Reference System, and the GeometricCompensation Reference System. An apparatus and method may providemultilevel programming, reading, and margining.

A sense amplifier system includes local sense amplifiers coupled tomemory subarrays and global sense amplifiers coupled to groups of localsense amplifiers.

Method and apparatus described herein are applicable to digitalmultilevel as well as analog multilevel system.

Memory Cell Technology

To facilitate the understanding of the invention, a brief description ofa memory cell technology is described below. In an embodiment theinvention applies to Source Side Injection (SSI) flash memory celltechnology, which will be referred to as SSI flash memory celltechnology. The invention is equally applicable to other technologiessuch as drain-side channel hot electron (CHE) programming (ETOX),P-channel hot electron programming, NROM (nitride programmable read onlymemory), SONOS (silicon-oxide-nitride-oxide-silicon), MONOS(metal-oxide-nitride-oxide-silicon), 2-D or 3-D flash, bi-directionalmemory cell (e.g., two storage nodes, one near drain and one near sourceof a memory cell; two floating gates of same one memory cell), phasechange memory, molecular memory, polymer memory, spin memory, singleelectron memory, nano particle memory, other hot electron programmingschemes, Fowler-Nordheim (FN) tunneling, ferro-electric memory, andother types of memory technology.

A cell structure of one typical SSI flash cell is symbolically shown inFIG. 1A. Its corresponding transistor symbol is shown in FIG. 1B. Thecell is made of two polysilicon gates (abbreviated as poly), a floatinggate poly FG 100F and a control gate poly CG 100C. The control gate CG100C also acts as a select gate that individually select each memorycell. This has the advantage of avoiding the over erase problem which istypical of stacked gate CHE flash cell. The floating gate has a poly tipstructure that points to the CG 100C, this is to enhance the electricfield from the FG 100F to the CG 100C which allows a much lower voltagein FN erase without using a thin interpoly oxide.

The thicker interpoly oxide leads to a higher reliability memory cell.The cell is also fabricated such that a major portion of the FG 100Foverlaps the source junction 100S. This is to make a very high couplingratio from the source 100S to FG 100F, which allows a lower erasevoltage and is advantageous to the SSI programming, which will bedescribed shortly. A structural gap between the FG 100F and at CG 100Cis also advantageous for the efficient SSI programming.

The SSI flash memory cell enables low voltage and low power performancedue to its intrinsic device physics resulting from its device structure.The SSI flash cell uses efficient FN tunneling for erase and efficientSSI for programming. The SSI flash cell programming requires a smallcurrent in hundreds of nano amps and a moderate voltage range of ˜8 to11 volts. This is in contrast to that of a typical drain-side channelhot electron memory cell programming which requires current in hundredsof microamp to milliamp range and a voltage in the range of 11 to 13volts.

The SSI flash memory cell erases by utilizing Fowler-Nordheim tunnelingfrom the floating gate poly to the control gate poly by applying a higherase voltage on the control gate CG 100C, e.g., 8–13 volts, and a lowvoltage on the source 100S, e.g., 0–0.5 volts. The high erase voltagetogether with high coupling from the source to the floating gate createsa localized high electric field from the FG 100F tip to the CG 100C andcauses electrons to tunnel from the FG 100F to the CG 100C near the tipregion. The resulting effect causes a net positive charge on the FG100F.

The SSI flash memory cell programs by applying a high voltage on thesource 100S (herein also known as common line CL), e.g., 4–13 V, a lowvoltage on the CG 100C, e.g., 0.7–2.5 V, and a low voltage on the drain100D (herein also known as the bitline BL), e.g., 0–1V. The high voltageon the source 100S strongly couples to the FG to strongly turn on thechannel under the FG (it will be equivalently referred to as the FGchannel). This in turn couples the high voltage on the source 100Stoward the gap region. The voltage on the CG 100C turns on the channeldirectly under the CG 100C (it will be equivalently referred to as theCG channel). This in turn couples the voltage on the drain 100D towardthe gap region. Hence, the electrons flow from the drain junction 100Dthrough the CG channel, through the gap channel, through the FG channel,and finally arrive at the source junction.

Due to the gap structure between the CG 100C and the FG 100F, in thechannel under the gap, there exists a strong lateral electric field(EGAPLAT) 100G. As the EGAPLAT 100G reaches a critical field, electronsflowing across the gap channel become hot electrons. A portion of thesehot electrons gains enough energy to cross the interface between thesilicon and silicon dioxide into the silicon dioxide. And as thevertical field Ev is very favorable for electrons to move from thechannel to the FG 100F, many of these hot electrons are swept toward theFG 100F, thus, reducing the voltage on the FG 100F. The reduced voltageon the FG 100F reduces electrons flowing into the FG 100F as programmingproceeds.

Due to the coincidence of favorable Ev and high EGAPLAT 100G in the gapregion, the SSI memory cell programming is more efficient over that ofthe drain-side CHE programming, which only favors one field over theother. Programming efficiency is measured by how many electrons flowinto the floating gate as a portion of the current flowing in thechannel. High programming efficiency allows reduced power consumptionand parallel programming of multiple cells in a page mode operation.

Multilevel Memory Integrated Circuit System

The challenges associated with putting together a billion transistors ona single chip without sacrificing performance or cost are tremendous.The challenges associated with designing consistent and reliablemultilevel performance for a billion transistors on a single chipwithout sacrificing performance or cost are significantly moredifficult. The approach taken here is based on the modularizationconcept. Basically everything begins with a manageable optimized basicunitary block. Putting appropriate optimized unitary blocks togethermakes the next bigger optimized block.

A super high density nonvolatile multilevel memory integrated circuitsystem herein described is used to achieve the performance targets ofread speed, write speed, and an operating lifetime with low cost. Readspeed refers to how fast data could be extracted from a multilevelmemory integrated circuit system and made available for external usesuch as for the system microcontroller 2001 shown in FIG. 1C which isdescribed later. Write speed refers to how fast external data could bewritten into a multilevel memory integrated circuit system. Operatinglifetime refers to how long a multilevel memory integrated circuitsystem could be used in the field reliably without losing data.

Speed is modularized based on the following concept, T=CV/I, whereswitching time T is proportional to capacitance C multiplied by thevoltage swing V divided by the operating current I. Methods andapparatuses are provided by the invention to optimize C, V, and I toachieve the required specifications of speed, power, and optimal cost toproduce a high performance high-density multilevel memory integratedcircuit system. The invention described herein makes the capacitanceindependent of memory integrated circuit density, to the first order,and uses the necessary operating voltages and currents in an optimalmanner.

A nonvolatile multilevel memory system is shown in FIG. 1C. A super highdensity nonvolatile multilevel memory integrated circuit (IC) system2000 is a digital multilevel nonvolatile flash memory integrated circuitcapable of storing 2^(N) storage levels per one memory cell, withN=number of digital bits. A system microcontroller 2001 is a typicalsystem controller used to control various system operations. Controlsignals (CONTROL SIGNALS) 196L, input/output bus (IO BUS) 194L, andready busy signal (R/BB) 196RB are for communication between the systemmicrocontroller 2001 and the super high density nonvolatile multilevelmemory integrated circuit system 2000.

An electronic camera system (SILICONCAM) 2008 utilizing super highdensity nonvolatile multilevel memory IC system 2000 is shown in FIG.1D. The system (SILICONCAM) 2008 includes an integrated circuit system(ECAM) 2005 and an optical lens block (LENS) 2004. The integratedcircuit system (ECAM) 2005 includes an image sensor (IMAGE SENSOR) 2003,an analog to digital converter block (A/D CONVERTER) 2002, a systemmicrocontroller 2001, and the multilevel memory IC system 2000. Theoptical lens block (LENS) 2004 is used to focus light into the IMAGESENSOR 2003, which converts light into an analog electrical signal. TheIMAGE SENSOR 2003 is a charge coupled device (CCD) or a CMOS sensor. Theblock (A/D CONVERTER) 2002 is used to digitize the analog electricalsignal into digital data. The microcontroller 2001 is used to controlvarious general functions such as system power up and down, exposuretime and auto focus. The microcontroller 2001 is also used to processimage algorithms such as noise reduction, white balance, imagesharpening, and image compression. The digital data is stored in themultilevel memory IC system 2000. The digital data can be down loaded toanother storage media through wired or wireless means. Future advancesin process and device technology can allow the optical block (LENS) 2004to be integrated in a single chip with the ECAM 2005.

An electronic audio system (SILICONCORDER) 2007 utilizing super highdensity nonvolatile multilevel memory IC system 2000 is shown in FIG.1E. The SILICONCORDER 2007 includes an integrated circuit system(SILICONAUDIO) 2006, a MICROPHONE 2012, and a SPEAKER 2013. The system(SILICONAUDIO) 2006 includes an anti-alias FILTER 2010, an A/D CONVERTER2002, a smoothing FILTER 2011, a D/A CONVERTER 2009, a systemmicrocontroller 2001, and the multilevel memory IC system 2000. TheFILTER 2010 and the FILTER 2011 can be combined into one filter block ifthe signals are multiplexed appropriately. The microcontroller 2001 isused to control various functions such as system power up and down,play, record, message management, audio data compression, and voicerecognition. In recording a sound wave, the MICROPHONE 2012 converts thesound wave into an analog electrical signal, which is filtered by theFILTER 2010 to reduce non-audio signals. The filtered analog signal isthen digitized by the A/D CONVERTER 2002 into digital data. The digitaldata is then stored in compressed or uncompressed form in the multilevelmemory IC system 2000. In playing back the stored audio signal, themicrocontroller 2001 first uncompresses the digital data if the data isin compressed form. The D/A CONVERTER 2009 then converts the digitaldata into an analog signal which is filtered by a smoothing filter(FILTER) 2011. The filtered output analog signal then goes to theSPEAKER 2013 to be converted into a sound wave. The signal filtering canbe done by digital filtering by the microcontroller 2001. Externaldigital data can be loaded into the multilevel memory IC system 2000through wired or wireless means. Future advances in process and devicetechnology can allow the MICROPHONE 2012 and the SPEAKER 2013 to beintegrated in a single chip with the SILICONAUDIO 2006.

A circuit block diagram of the super high density nonvolatile multilevelmemory integrated circuit system 2000 based on the concepts describedabove and also on ideas described below, is shown in FIG. 2A. For thepurpose of discussion, a giga bit nonvolatile multilevel memory chip isdescribed.

A circuit block 100 includes a regular memory array.

It includes a total of for example, 256 million nonvolatile memory cellsfor a 4-bit digital multilevel memory cell technology or 128 millioncells for a 8-bit digital multilevel memory cell technology. An N-bitdigital multilevel cell is defined as a memory cell capable of storing2^(N) levels. A reference array (MFLASHREF) 106 is used for thereference system. A redundancy array (MFLASHRED) 102 is used to increaseproduction yield by replacing bad portions of the regular memory arrayof the circuit block 100. An optional spare array (MFLASHSPARE) 104 canbe used for extra data overhead storage such as for error correction.

A y-driver block (YDRV) 110 including a plurality of single y-drivers(YDRVS) 110S is used for controlling the bitlines during write, read,and erase operation. Block YDRVS 110S will be described in detail belowin the description of the multilevel algorithm. Multiples of y-driverblock (YDRV) 110 are used for parallel multilevel page writing andreading to speed up the data rate during write to and read from themultilevel memory IC system 2000. A reference y-driver block (REFYDRV)116 including a plurality of single reference y-drivers (REFYDRVS) 116Sis used for the reference array block (MFLASHREF) 106. A redundanty-driver block (RYDRV) 112 including a plurality of single redundanty-drivers (RYDRVS) 112S is used for the redundant array (MFLASHRED) 102.The function of block (RYDRVS) 112S is similar to that of block (YDRVS)110S. A spare y-driver block (SYDRV) 114 including a plurality of singlespare y-drivers (SYDRVS) 114S is used for the spare array (MFLASHSPARE)104. The function of block (SYDRVS) 114S is similar to that of block(YDRVS) 110S. A page select block (PSEL) 120 is used to select onebitline out of multiple bitlines for each single y-driver (YDRVS) 110Sinside the block (YDRV) 110. Corresponding select circuit blocks forreference array, redundant array, and spare array are a reference pageselect block (PRSEL) 126, a redundant page select block 122, and a sparepage select block 124. A byte select block (BYTESEL) 140 is used toenable one byte data in or one byte data out of the blocks (YDRV) 110 ata time. Corresponding blocks for reference array, redundant array, andspare array are a reference byte select block 146, a redundant byteselect block 142, and a spare byte select block 144. The control signalsfor circuit blocks 116, 126, 146, 112, 122, 142, 114, 124, and 144 arein general different from the control signals for circuit blocks 110,120, and 140 of the regular memory array of the circuit block 100. Thecontrol signals are not shown in the figures.

A multilevel memory precision decoder block (MLMDEC) 130 is used foraddress selection and to provide precise multilevel bias levels overtemperature, process corners, and power supply as required forconsistent multilevel memory operation for the regular memory array ofthe circuit block 100 and for the redundant array 102. A multilevelmemory precision decoder block (MLMSDEC) 134 is used for addressselection and to provide precise multilevel bias levels overtemperature, process corners, and power supply as required forconsistent multilevel memory operation for the spare array 104.

An address pre-decoding circuit block (XPREDEC) 154 is used to providedecoding of addresses A<16:AN>. The term AN denotes the most significantbit of addresses depending on the size of the memory array. The outputsof block (XPREDEC) 154 couple to blocks (MLMDEC) 130 and block (MLMSDEC)134. An address pre-decoding block (XCGCLPRED) 156 is used to providedecoding of addresses A<11:15>. The outputs of block 156 also couple toblocks (MLMDEC) 130 and block (MLMSDEC) 134.

A page address decoding block (PGDEC) 150 is used to provide decoding ofaddresses A<9:10>. The outputs of block (PGDEC) 150 couple to blocks(PSEL) 120. A byte address decoding block (BYTEDEC) 152 is used toprovide decoding of addresses A<0:8>. The outputs of block (BYTEDEC) 152couple to blocks (BYTESEL) 140. An address counter block (ADDRCTR) 162provides addresses A<11:AN>, A<9:10>, and A<0:8>for row, page, and byteaddresses, respectively. The outputs of the block (ADDRCTR) 162 coupleto blocks (XPREDEC) 154, (XCGCLPRED) 156, (PGDEC) 150, and (BYTEDEC)152. The inputs of the block (ADDRCTR) 162 are coupled from the outputsof an input interface logic block (INPUTLOGIC) 160.

The input interface logic block (INPUTLOGIC) 160 is used to provideexternal interface to systems off-chip such as the microcontroller 2001.Typical external interface for memory operation are read, write, erase,status read, identification (ID) read, ready busy status, reset, andother general purpose tasks. Serial interface can be used for the inputinterface to reduce pin counts for high-density chip due to a largenumber of addresses. Control signals 196L are used to couple theINPUTLOGIC 160 to the system microcontroller 2001. The INPUTLOGIC 160includes a status register that is indicative of the status of thememory chip operation such as pass or fail in program or erase, ready orbusy, write protected or unprotected, cell margin good or bad, restoreor no restore, etc. The margin and restore concepts are described morein detail in the multilevel algorithm description.

An algorithm controller block (ALGOCNTRL) 164 is used to handshake theinput commands from the block (INPUTLOGIC) 160 and to execute themultilevel erase, programming and sensing algorithms as needed formultilevel nonvolatile operation. The ALGOCNTRL 164 is also used toalgorithmically control the precise bias and timing conditions asrequired for multilevel precision programming.

A test logic block (TESTLOGIC) 180 is used to test various electricalfeatures of the digital circuits, analog circuits, memory circuits, highvoltage circuits, and memory array. The inputs of the block (TESTLOGIC)180 are coupled from the outputs of the INPUTLOGIC 160. The block(TESTLOGIC) 180 also provides timing speed-up in production testing suchas faster write/read and mass modes. The TESTLOGIC 180 is also used toprovide screening tests associated with memory technology such asvarious disturb and reliability tests. The TESTLOGIC 180 also allows anoff-chip memory tester to directly take over the control of variouson-chip logic and circuit bias blocks to provide various externalvoltages and currents and external timing. This feature permits, forexample, screening with external voltage and external timing or permitsaccelerated production testing with fast external timing.

A fuse circuit block (FUSECKT) 182 is a set of nonvolatile memory cellsconfigured at the external system level, at the tester, at the user, oron chip on-the-fly to achieve various settings. These settings caninclude precision bias levels, precision on-chip oscillator,programmable logic features such as write-lockout feature for portionsof an array, redundancy fuses, multilevel erase, program and readalgorithm parameters, or chip performance parameters such as write orread speed and accuracy.

A reference control circuit block (REFCNTRL) 184 is used to provideprecision reference levels for precision voltage levels as required formultilevel programming and sensing.

A redundancy controller block (REDCNTRL) 186 is for redundancy controllogic.

A voltage algorithm controller block (VALGGEN) 176 provides variousspecifically shaped voltage signals of amplitude and duration asrequired for multilevel nonvolatile operation and to provide precisevoltage levels with tight tolerance, as required for precisionmultilevel programming, erasing, and sensing.

A circuit block (BGAP) 170 is a bandgap voltage generator based on thebandgap circuit principle to provide a precise voltage level overprocess, temperature, and supply as required for multilevel programmingand sensing.

A voltage and current bias generator block (V&IREF) 172 is an on-chipprogrammable bias generator. The bias levels are programmable by thesettings of the control signals from the FUSECKT 182 and also by variousmetal options. A precision oscillator block (PRECISIONOSC) 174 providesaccurate timing as required for multilevel programming and sensing.

Input buffer blocks 196 are typical input buffer circuits, for example,TTL input buffers or CMOS input buffers. Input/output (io) buffer blocks194 includes typical input buffers and typical output buffers. A typicaloutput buffer is, for example, an output buffer with slew rate control,or an output buffer with level feedback control. A circuit block 196R isan open drained output buffer and is used for ready busy handshakesignal (R/BB) 196RB.

A voltage multiplier (also known as charge pump) block (VMULCKT) 190provides voltage levels above the external power supply required forerase, program, read, and production tests. A voltage multiplyingregulator block (VMULREG) 192 provides regulation for the block(VMULCKT) 190 for power efficiency and for transistor reliability suchas to avoid various breakdown mechanisms.

A flash power management block (FPMU) 198 is used to efficiently managepower on-chip such as powering up only the circuit blocks in use. TheFPMU 198 also provides isolation between sensitive circuit blocks fromthe less sensitive circuit blocks by using different regulators fordigital power (VDDD) 1032/(VSSD) 1033, analog power (VDDA) 1030/(VSSA)1031, and IO buffer power (VDDIO) 1034/(VSSIO) 1035. The FPMU 198 alsoprovides better process reliability by stepping down power supply VDD tolower levels required by transistor oxide thickness. The FPMU 198 allowsthe regulation to be optimized for each circuit type. For example, anopen loop regulation could be used for digital power since highlyaccurate regulation is not required; and a closed loop regulation couldbe used for analog power since analog precision is normally required.The flash power management also enables creation of a “green” memorysystem since power is efficiently managed.

Block diagram of the FPMU 198 is shown in FIG. 2B. A VDD 1111 and a VSS1000 are externally applied power supply and ground lines, respectively.A block (ANALOG POWER REGULATOR) 198A is an analog power supplyregulator, which uses closed loop regulation. The closed loop regulationis provided by negative feedback action of an operational amplifier (opamp) 1003 configured in a voltage buffer mode with a reference voltage(VREF1) 1002 on the positive input of the op amp 1003. A filtercapacitor (CFILL) 1004 is used for smoothing transient response of theanalog power (VDDA) 1030. A ground line (VSSA) 1031 is for analog powersupply. A block (DIGITAL POWER REGULATOR) 198B is a digital power supplyregulator, which uses open loop regulation. The open loop regulation isprovided by source follower action of a transistor 1006 with a referencevoltage (VREF2) 1005 on its gate. A pair of filter capacitor (CFIL4)1009 and (CFIL2) 1007 are used for smoothing transient response ofdigital power (VDDD) 1032. A loading element (LOAD1) 1008 is for thetransistor 1006. A ground line (VSSD) 1033 is for digital power supply.A block (IO POWER REGULATOR) 198C is an IO power supply regulator, whichuses open loop regulation similar to that of the digital power supply198B. The open loop regulation is provided by a transistor 1011 with areference voltage (VREF3) 1010 on its gate. A loading element (LOAD2)1013 is for transistor 1011. A pair of capacitors (CFIL5) 1014 and(CFIL3) 1012 are used for smoothing transient response of IO power(VDDIO) 1034. A ground line (VSSIO) 1035 is for IO power supply. A block198D includes various circuits that require unregulated power supplysuch as transmission switches, high voltage circuits, ESD structures,and the like.

A block (PORK) 1040 is a power on reset circuit which provides a logicsignal (PON) 1041 indicating that the power supply being applied to thechip is higher than a certain voltage. The signal (PON) 1041 istypically used to initialize logic circuits before chip operationbegins.

A block (VDDDET) 1050 is a power supply detection circuit, whichprovides a logic signal (VDDON) 1051 indicating that the operating powersupply is higher than a certain voltage. The block (VDDDET) 1050 isnormally used to detect whether the power supply is stable to allow thechip to take certain actions such as stopping the programming if thepower supply is too low.

A block (FPMUCNTRL) 1060 is a power supply logic controller, thatreceives control signals from blocks (PORK) 104, (VDDDET) 1050,(INPUTLOGIC) 160, (ALGOCNTRL) 164, and other logic control blocks topower up and power down appropriately power supplies and circuit blocks.The FPMUCNTRL 1060 is also used to reduce the power drive ability ofappropriate circuit blocks to save power. A line (PDDEEP) 1021 is usedto power down all regulators. Lines (PDAPOW) 1020, (PDDPOW) 1022, and(PDIOPOW) 1023 are used to power down blocks 198A, 198B, and 198C,respectively. Lines (Pt)DEEP) 1021, (PDAPOW) 1020, (PDDPOW) 1022, and(PDIOPOW) 1023 come from block (FPMUCNTRL) 1060.

It is possible that either closed or open loop regulation could be usedfor any type of power supply regulation. It is also possible that anypower supply could couple directly to the applied power supply (VDD)1111 without any regulation with appropriate consideration. For example,VDDA 1030 or VDDIO 1034 could couple directly to VDD 1111 if highvoltage transistors with thick enough oxide are used for analog circuitsor IO buffer circuits, respectively.

A typical memory system operation is as follows: a host such as themicrocontroller 2001 sends an instruction, also referred to as acommand, such as a program instruction via the CONTROL SIGNALS 196L andthe IO BUS 194L to the multilevel memory chip 2000 (see FIG. 1C). TheINPUTLOGIC 160 interprets the incoming command as a valid command andinitiates the program operation internally. The ALGOCNTRL 164 receivesthe instruction from the INPUTLOGIC 160 to initiate the multilevelprogramming algorithmic action by outputting various control signals forthe chip. A handshake signal such as the ready busy signal R/BB 196RBthen signals to the microcontroller 2001 that the multilevel memory chip2000 is internally operating. The microcontroller 2001 is now free to doother tasks until the handshake signal R/BB 196RB signals again that themultilevel memory chip 2000 is ready to receive the next command. Atimeout could also be specified to allow the microcontroller 2001 tosend the commands in appropriate times.

Read Operation

A read command including a read operational code and addresses is sentby the microcontroller 2001 via the CONTROL SIGNALS 196L and IO BUS194L. The INPUTLOGIC 160 decodes and validates the read command. If itis valid, then incoming addresses are latched in the ADDRCTR 162. Theready busy signal (R/BB) 196RB now goes low to indicate that themultilevel memory device 2000 has begun read operation internally. Theoutputs of ADDRCTR 162 couple to blocks (XPREDEC) 154, (XCGCLPRED) 156,(PGDEC) 150, (BYTEDEC) 152, and (REDCNTRL) 186. The outputs of blocks154, 156, 150, 152, and 186 couple to blocks (MLMDEC) 130, (MLSMDEC)134, and block 100 to enable appropriate memory cells. Then theALGOCNTRL 164 executes a read algorithm. The read algorithm will bedescribed in detail later in the multilevel algorithm description. Theread algorithm enables blocks (BGAP) 170, (V&IREF) 172, (PRECISIONOSC)174, (VALGGEN) 176, and (REFCNTRL) 184 to output various precisionshaped voltage and current bias levels and algorithmic read timing forread operation, which will be described in detail later in thedescription of the multilevel array architecture. The precision biaslevels are coupled to the memory cells through blocks (MLMDEC) 130,(MLMSDEC) 134, and block 100.

In an embodiment, the read algorithm operates upon one selected page ofmemory cells at a time to speed up the read data rate. A page includes aplurality of memory cells, e.g., 1024 cells. The number of memory cellswithin a page can be made programmable by fuses, e.g., 512 or 1024 tooptimize power consumption and data rate. Blocks (PGDEC) 150, (MLMDEC)130, (MLMSDEC) 134, 100, and (PSEL) 120 select a page. All memory cellsin the selected page are put in read operating bias condition throughblocks (MLMDEC) 130, (MLMSDEC) 134, 100, (PSEL) 120, and (XCGCLPRED)156. After the readout voltage levels are stable, a read transfer cycleis initiated by the block (ALGOCNTRL) 164. All the readout voltages fromthe memory cells in the selected page are then available at they-drivers (YDRVS) 110S, (RYDRVS) 112S, and (SYDRVS) 114S inside block(YDRV) 110, (RYDRV) 112, and (SYDRV) 114, respectively.

Next, in the read transfer cycle the ALGOCNTR 164 executes a multilevelread algorithm to extract the binary data out of the multilevel cellsand latches them inside the YDRVS 110S, RYDRVS 112S, and SYDRVS 114S.This finishes the read transfer cycle. A restore flag is now set orreset in the status register inside the INPUTLOGIC 160. The restore flagindicates whether the voltage levels of the multilevel memory cellsbeing read have been changed and whether they need to be restored to theoriginal voltage levels. The restore concept will be described more indetail in the multilevel algorithm description. Now the ready busysignal (R/BB) 196RB goes high to indicate that the internal readoperation is completed and the multilevel memory device 2000 is ready totransfer out the data or chip status. The microcontroller 2001 now canexecute a status read command to monitor the restore flag or execute adata out sequence. The data out sequence begins with an external readdata clock provided by the microcontroller 2001 via the CONTROL SIGNAL196L coupled to an input buffer 196 to transfer the data out. Theexternal read data clock couples to the blocks (BYTEDEC) 152 and(BYTESEL) 140, 142, and 144 to enable the outputs of the latches insideblocks (YDRV) 110 or (RYDRV) 112 or (SYDRV) 114 to output one byte ofdata at a time into the bus 10<0:7> 1001. The external read data clockkeeps clocking until all the desired bytes of the selected page areoutputted. The data on bus IO<0:7> 1001 is coupled to themicrocontroller 2001 via IO BUS 194L through IO buffers 194.

Program Operation

A program command including a program operational code, addresses, anddata is sent by the microcontroller 2001 via CONTROL SIGNALS 196L and IOBUS 194L. The INPUTLOGIC 160 decodes and validates the command. If it isvalid, then incoming addresses are latched in the ADDRCTR 162. The datais latched in the latches inside YDRV 110, RYDRV 112, and SYDRV 114 viablocks (BYTEDEC) 152, (BYTESEL) 140, 142, and 144, respectively. Theready busy signal (R/BB) 196RB now goes low to indicate that the memorydevice has begun program operation internally. The outputs of ADDRCTR162 couple to blocks (XPREDEC) 154, (XCGCLPRED) 156, (PGDEC) 150,(BYTEDEC) 152, and (REDCNTRL) 186. The outputs of blocks 154, 156, 150,152, and 186 couple to blocks (MLMDEC) 130, (MLSMDEC) 134, and 100 toenable appropriate memory cells. Then the (ALGOCNTRL) 164 executes aprogram algorithm, which will be described in detail later in themultilevel algorithm description. The (ALGOCNTRL) 164 enables blocks(BGAP) 170, (V&IREF) 172, (PRECISIONOSC) 174, (VALGGEN) 176, and(REFCNTRL) 184 to output various precision shaped voltage and currentbias levels and algorithmic program timing for the program operation,which will be described in detail later in the description of themultilevel array architecture. The precision bias levels are coupled tothe memory cells through blocks (MLMDEC) 130, (MLMSDEC) 134, and block100.

In an embodiment, the program algorithm operates upon one selected pageof memory cells at a time to speed up the program data rate. Blocks(PGDEC) 150, (MLMDEC) 130, (MLMSDEC) 134, 100, and (PSEL) 120 select apage. All memory cells in the selected page are put in appropriateprogram operating bias condition through blocks (MLMDEC) 130, (MLMSDEC)134, 100, (PSEL) 120, and (XCGCLPRED) 156. Once the program algorithmfinishes, program flags are set in the status register inside the block(INPUTLOGIC) 160 to indicate whether the program has been successful.That is, all the cells in the selected page have been programmedcorrectly without failure and with enough voltage margins. The programflags are described more in detail in the multilevel algorithmdescription. Now the ready busy signal (R/BB) 196RB goes high toindicate that the internal program operation is completed and the memorydevice is ready to receive the next command.

Erase Operation

An erase command including an erase operational code and addresses issent by the microcontroller 2001 via CONTROL SIGNALS 196L and IO BUS194L. The INPUTLOGIC 160 decodes and validates the command. If it isvalid, then incoming addresses are latched in the ADDRCTR 162. The readybusy signal (R/BB) 196RB now goes low to indicate that the memory devicehas begun erase operation internally. The outputs of ADDRCTR 162 coupleto blocks (XPREDEC) 154, (XCGCLPRED) 156, (PGDEC) 150, (BYTEDEC) 152,and (REDCNTRL) 186. The outputs of blocks 154, 156, 150, 152, and 186couple to blocks (MLMDEC) 130, (MLSMDEC) 134, and 100 to enableappropriate memory cells. Then the ALGOCNTRL 164 executes an erasealgorithm. The ALGOCNTRL 164 enables blocks (BGAP) 170, (V&IREF) 172,(PRECISIONOSC) 174, (VALGGEN) 176, and (REFCNTRL) 184 to output variousprecision shaped voltage and current bias levels and algorithmic erasetiming for erase operation. The shaped voltage for erase is to minimizeelectric field coupled to memory cells, which minimizes the damage tomemory cells during erasing. The precision bias levels are coupled tothe memory cells through blocks (MLMDEC) 130, (MLMSDEC) 134, and block100.

In an embodiment, the erase algorithm operates upon one selected eraseblock of memory cells at a time to speed up the erase time. An eraseblock includes a plurality of pages of memory cells, e.g., 32 pages. Thenumber of pages within an erase block can be made programmable by fusesto suit different user requirements and applications. Blocks (PGDEC)150, (MLMDEC) 130, (MLMSDEC) 134, 100, and (PSEL) 120 select a block.All memory cells in the selected block are put in erase operating biascondition through blocks (MLMDEC) 130, (MLMSDEC) 134, 100, (PSEL) 120,and (XCGCLPRED) 156. Once the erase algorithm finishes, the erase flagsare set in the status register inside the block (INPUTLOGIC) 160 toindicate whether the erase has been successful. That is, all the cellsin the selected page have been erased correctly to desired voltagelevels without failure and with enough voltage margins. Now the readybusy signal (RIBB) 196RB goes high to indicate that the internal eraseoperation is completed and the multilevel memory device 2000 is ready toreceive the next command.

Multilevel Array Architecture

The demanding requirements associated with putting together a billiontransistors on a single chip with the ability to store multipleprecision levels per cell and operating at a very high speed arecontradictory. These requirements need innovative approaches and carefultradeoffs to achieve the objective. Examples of tradeoffs and problemswith prior art implementation are discussed below. In conventional priorart architectures, a voltage drop along a metal line of a few tens ofmillivolts could be easily tolerated. Here, in a super high densitynonvolatile multilevel memory integrated circuit system such a voltagedrop can cause unacceptable performance degradation in precision levelsdue to the high number of levels stored per memory cell. In conventionalarray architectures, a bit line capacitance in the order of 10 picofarads would be a non-issue. Here it may be unworkable due to the highdata rate required. In prior art array architectures a bias levelvariation from one memory cell to another in the order of +/−30 percentwould be a typical situation. Here such a bias variation would be aserious performance problem. In prior art array architectures, the totalresistance of a memory source line in the order of a few hundreds ofohms would be a typical situation, here a few tens of ohms is a seriousproblem. The huge number of memory cells of the giga to tera bithigh-density memory system compounds the matter even further by makingthe memory source line longer. Another challenge facing the multilevelsystem is maintaining high speed sensing and programming with low power,again requiring tradeoffs. Another challenge facing the multilevelsystem is high speed sensing and programming with very high precisionvoltages due to a high number of levels stored per digital multilevelmemory cell, again a conflicting demand. Another challenge facing themultilevel system is high speed sensing and programming consistentlyevery time over many years, process corners, temperature, and powersupply variation.

To get an appreciation of the order of magnitude of the difficultyinvolved in the super high density multilevel nonvolatile memory system,numerical examples will be given corresponding to a one giga bit arrayarchitecture system suitable for 256 levels, i.e., 8 bits. The array isthen organized as 8192 bitlines or columns and 16384 rows or wordlinesfor a total of 134,217,730 physical cells.

One sensing level, V1level,=multilevel sensing range/2^(N), N=number ofdigital bits stored per memory cell. Multilevel sensing range is thereadout voltage range from sensing a multilevel memory cell. Assumingthe multilevel sensing range from the multilevel memory cell availableis 2048 millivolts, then V1level=2048/256=8 millivolts.

A very high data rate is required for applications such as image or highdensity data storage. For example, write and read rates of a mega byteper second are required. To achieve this high data rate, parallelwriting and sensing is required for the super high density nonvolatilemultilevel memory integrated circuit system. In the present embodiment,a total of 1024 y-drivers (YDRVS) 110S inside blocks (YDRV) 110 areused. This allows 1024 memory cells to be written and sensed at the sametime in a page mode manner, effectively increasing the speed by a factorof 1024 over single cell operation. The number of bitlines multiplexedinto one single y-driver (YDRVS) 110S is=8192/1024=8 bitlines.

A program algorithm described in more detail elsewhere in thisspecification is able to achieve desired multilevel resolution. The reador program multilevel resolution is the smallest voltage range in reador program, respectively, needed to operate the multilevel memory cellscorrectly. An erase algorithm first erases the memory cells to make thecell readout voltage reaching a certain desired voltage level. Then theiterative program algorithm is applied to the memory cells. The programalgorithm includes a plurality of verify-program cycles. Averify-program cycle includes a verify cycle followed by a programcycle. A verify cycle is done first to inhibit the cell from the firstprogramming pulse if the cell is verified, therefore preventing possibleover-programming. Over-programming means that after a programming pulsethe cell sensing level passes a desired voltage level by more than adesired voltage amount. A verify cycle is used to determine whether thedesired readout sensing level has been reached. If the desired readoutsensing level is reached, the cell is inhibited from furtherprogramming. Otherwise, the cell is enabled for the next program cycle.A program cycle is used to change incrementally the charge stored in thecell and the corresponding cell sensing readout voltage. Instead of averify-program cycle, a program-verify cycle can be used. Aprogram-verify cycle begins with a program cycle followed by a verifycycle. In this case, care should be taken to ensure that the firstprogramming pulse does not cause over-programming.

In an embodiment the program cycle includes applying a voltage on thesource line, (interchangeably referred to as common line [CL]) (VCL),with a predetermined program pulsewidth (TPPWD) and a predeterminedprogram bias cell current (Ipcell). The verify cycle makes use of thevoltage mode sensing as shown in FIG. 2C, which applies a referencevoltage (VCLRD) on the source line (CL), another reference voltage(VCGRD) on the control gate, and a predetermined read bias current(Ircell) on the bitline and through the memory cell. The current(Ircell) is applied to the bitline and the memory cell through selecttransistors which are not shown. The resulting voltage on the bitline isthe sensing readout voltage (VR), which has a unique relationship to thecharge on the floating gate. The voltage mode sensing is also usedduring read. In another embodiment of voltage mode sensing, the sourceline (CL) and the bitline are interchanged, and thus a reference voltageis applied on the bitline and a predetermined read bias current (Ircell)is applied on the source line through the memory cell. The resultingvoltage on the source line is the sensing readout voltage (VR). In thiscase, the array architecture uses only one source line in read at agiven time, for example, by multiplexing through decoder circuitry orover time. This is to be known as Inverse Voltage Mode sensing. Inanother embodiment of the voltage mode sensing, there is nopredetermined read current (Ircell), or the predetermined read currentequals to zero. This mode is to be known as No Current (Digital)Multilevel Mode Sensing. In another embodiment of the voltage modesensing, the predetermined read bias current is replaced by a resistoror an equivalent resistance (like a MOS operated as a resistor). Tochange incrementally the readout sensing voltage to the next value(VR+dVR), with dVR equals to the incremental readout sensing voltagechange, the next program cycle is repeated with the common line voltageincreased incrementally to (VCL+dVCLP), with dVCLP equals to theincremental programming voltage change.

The number of verify-program cycles (NC) is dependent on the number ofvoltage levels and various margins of the memory system. For example,for an equivalent 8-bit digital multilevel cell, there are 2^(N)=2⁸=256levels, with N=8. The minimum possible number of verify-program cycles(NC) required would be 256. To cover variations due to cell-to-cellvariation, temperature, process corners, an algorithm may require, forexample, approximately 1.4×256=360 verify-program cycles. To covervarious margins needed such as for data retention and programmingdistribution, the number of verify-program cycles required is actuallyhigher. Assuming a factor of 2 due to various margin coverage, thenumber of verify-program cycles is approximately equal to 720. The exactnumber of verify-program cycles is typically varied depending on variousmemory technologies and particular desired performance targets.

For write data rate of 1 mega byte per second and for 8-bit digitalmultilevel operation with 1024 bytes per page, the write timing per pageis, TWRT=# of bytes written in parallel/data rate=1024 bytes per page/1mega bytes/second=1024 μs=1.024 ms per page.

Hence the time to execute each program-verify cycle (TPV) must be lessthan TWRT/NC=1.024 ms/720=1.42 μs. This fast timing coupled withparallel operation of 1024 cells has important implication on memorycell program speed, capacitance loading, power consumption and othereffects as will be described below.

Typical process parameters of a sub-micron memory cell are as follows. Atypical diffused source line resistance per cell is 100 ohms. A typicalbitline resistance per cell is 80 milliohms. A typical silicided rowline resistance per cell is 20 ohms. A typical source line capacitanceper cell is 2 fF. A typical bitline capacitance per cell is 1.5 fF. Anda typical row line capacitance per cell is 3 fF.

Hence for the 8192×16384 array, the total bitline capacitance isCBL=˜16384×1.5 fF=25 pF, where “=˜” is defined as approximately equalto. The total metal bitline resistance RBL=˜16384×0.08=1330 ohms. Thetotal diffused source line resistance is RSL=8192×100=819 K ohms. Thetotal row line resistance is RWL=8192×20=164 K ohms. For a typicalmemory system, the diffused source line is strapped by metal along thesource line, with approximately 80 milliohms per cell, in this caseRSL=8192×0.08=655 ohms.

In conventional stacked gate drain-side CHE programming (abbreviated asCHE flash program), the single cell current is typically 1 ma, whichcauses a voltage drop along a single metal bitline of =˜1 ma×RBL=˜1ma×1330 ohms=1330 millivolts, which is unacceptable since it is muchgreater than 1 level=8 millivolts. In SSI flash programming (abbreviatedas SSI flash program), the typical cell current can be lowered to 1 μa,which causes a voltage drop along a single metal bitline of =˜1 μa×1330ohms=1.33 millivolts, which is acceptable.

For 1024 cells drawing the cell current (Icell) continuously, thevoltage drop (DVCL) along the source line from the driver to the otherend follows the geometric equation:DVCL=0.5*P*(P+1)*R8cell*Icell,tm  (1)where R8cell=the metal source line resistance for 8 cells in series=0.08ohms×8=0.64 ohms, and P=1024.

Along the source line, for 1024 cells programming simultaneously, thetotal current is 1024×1 ma=1.024 A for the CHE flash program and=1024×1μa=1.024 ma for the SSI flash program. The power needed for the drainside CHE flash programming for parallel page mode operation isunsustainable due to very high current. Additionally, the voltage dropalong the metal source line by equation (1) is =˜0.5×1024*1025*0.64*1ma=336 Volts for CHE. This is unworkable for CHE flash technology.Similarly, the source line voltage drop for the SSI flash =˜336millivolts. This is also unworkable in the multilevel program for thefollowing reasons.

For a multilevel nonvolatile system, in one program cycle, the cellsensing voltage can only shift (dVR) a maximum of <(Q*V1 level) forreliable sensing, where Q was 0.5 in the prior example. However Q couldvary from ⅓ to ⅛ for long term reliability. This is needed, for example,to allow for sensing margin, verify margin, program disturb, dataretention, and endurance. The number of cells programming simultaneouslywithin a selected page can vary between as many as 1024 to as few asonly one from one program cycle to the next. Thus the total programcurrent flowing through the common line CL could change by a factor of1024 from one program cycle to the next. The resulting worst casevoltage change in the source line VCL from, one program cycle to thenext is dVCL=˜336 millivolts for SSI flash. This voltage jump in VCLcauses the only remaining programming cell to over program, which causesthe cell sensing voltage to shift much greater than the (Q*V1 level).Hence, the challenge is to bring the voltage drop dVCL to an acceptablelevel during programming.

For verifying after programming multilevel memory cells, conventionalmethods would shut off the read cell currents for cells that havealready reached their desired verifying levels, this would cause thevoltage shift dVCL in verify as much as in programming as describedabove. This voltage jump dVCL would couple to the memory cells and causea large jump in cell sensing voltage. This undesired large jump in cellsensing voltage causes an error in sensing, herein called a sense errorVRerr. This sense error should be much less than (Q*V1 level). Hencethis large jump is unacceptable. The invention solves the problem byenabling the total current all the time whether the cells have beenverified or not. This mitigates the change in the source line voltage.However a new problem surfaces as compared to that in programming. Astemperature changes from −45 C. to +85 C. the resistance of the sourceline metal line changes by about 40%, hence the source line voltage dropchanges by about 40%, which causes an additional sense error VRerr inread. This sense error should be much less than (Q*V1 level) to preventoverall read margin degradation. Therefore, an array architecture isneeded to achieve this, as will be described in detail below.

With 1024 cells operating simultaneously, assuming sense currentIrcell=10 μa, the total sense current is=1024×10 μa=10.24 ma flowinginto the source line. This presents several problems. With powerspecification for a typical memory chip ICC=20–30 ma. This 10.24 ma is abig percentage of the power specification. To deliver 10.24 ma whilemaintaining a precise voltage level VCLRD, VCLRD is defined as thevoltage in read on CL line, requires a challenging decoding and driverscheme, which will be addressed in the description of the multileveldecoding scheme. Large current flowing across the source line alsocauses the voltage drop as described above.

High data rate, meaning high sense speed and write speed, is requiredfor data intensive application. The speed is proportional to capacitanceand voltage swing and inversely proportional to the current,T=C*V/I  (2).

For typical bitline capacitance as calculated above, CBL=25 pF andassuming voltage swing V=1V, and assuming available current I=10 μa, thetime it takes to charge or discharge a bitline as needed in verify orprogram cycle is, TBL=25 pF*1V/10 μa=2.5 μs. This is greater than theTPV=1.42 μs as calculated above. At least a 2× or better timing isrequired for TBL to allow for various settling time, sensing time, andprogramming time. Increasing the current would cause higher powerconsumption, large decoding driver, and voltage problems as describedabove.

Further, in programming 1024 cells in parallel, the programming currentis supplied from an on-chip voltage multiplier, also known as a chargepump. The on-chip voltage multiplier multiplies the low voltage powersupply, e.g., 2.5 V to the required higher voltages. Allowing areasonable area penalty from the on-chip voltage multiplier, a totalcurrent of 100 μa is allowed for programming. The programming currentper cell is 100 μa/1024=0.1 μa. This causes a TBL=25 pF*1V/0.1 μa=250μs, which is even more severe of a timing problem. Here an improvementof more than 2 order of magnitude or better in speed is needed. Theinvention describes array architectures with suitable operating methodsto achieve this improvement and will be described below.

FIG. 3A is the block diagram of a super high-density digital nonvolatilemultilevel memory array architecture which is capable of >8-bitmultilevel operation. The block 100 has been expanded from FIG. 2A toshow the sub-blocks inside. A multilevel precision memory decoderMLMDECS 132 is used for delivering bias voltage levels with tighttolerance over temperature, process, and power supply variation formultilevel memory cells. A multilevel memory sub-array MFLSUBARY 101includes a plurality of single multilevel memory cells. Other blocks inFIG. 3A have already been described in association with the descriptionof FIG. 2A.

A block (PSEL) 120 includes a plurality of circuit blocks (PSELS) 120S.FIG. 3B shows details of a page select circuit (PSELS) 120S that selectsa pair of bitlines at a time. Transistors 120A–D are select transistors.Transistors 120E-H are inhibit transistors. Lines (PP0) 120K, (PP1)120M, (PP2) 120O, and (PP3) 120Q are complementary signals of lines(PPOB) 120L, (PP1B) 120N, (PP2B) 120P, and (PP3B) 120R, respectively.Line (BLYDRV) 120Y goes to one y-driver (YDRVS) 110S inside the block(YDRV) 110. Block (YDRVS) 110S will be described in detail later in thedescription of the multilevel algorithm. Lines (BLTP0) 240P, (BLTP1)241P, (B LTP2) 242P, and (BLTP3) 243P couple to the bitlines in block101 and couple to a set of lines (BLP0) 240, (BLP1) 241, (BLP2) 242, and(BLP3) 243 of the circuit block 290 in FIG. 4A.

FIG. 3C shows a block diagram of a block (MFLSUBARY) 101. A block(MFLSUBARY) 101 includes a plurality of blocks (ARYSEG0) 290. Blocks(ARYSEG0) 290 are first tiled horizontally NH times and then thehorizontally tiled blocks 290 are tiled vertically NV times. For a pagewith 1024 memory cells, NH is equal to 1024. NV is determined such thatthe total number of memory cells is equal to the size of the desiredphysical memory array.

FIG. 4A shows a basic array unit (ARYSEG0) 290. A block (RD1SEG) 300 isa multilevel decoding block. A plurality of the blocks RDLSEG makes upthe circuit block (MLMDEC) 130. In the block (ARYSEGO) 290, there are 8columns and FIG. 4A shows only 8 rows of memory cells, while other rows,e.g., 120 rows, are not shown for clarity. Each ARYSEG0 290 includes aplurality, e.g. 8, of array blocks (ARYLBLK) 290A tiled vertically. Aset of transistors 220, 221, 222, 223, 224, 225, 226, 227 couplesrespectively a set of segment bitlines (SBL0) 240A and (SBL1) 240B,(SBL2) 241A and (SBL3) 241B, (SBL4) 242A and (SBL5) 242B, (SBL6) 243Aand (SBL7) 243B to a set of top bitlines (BLP0) 240, (BLP1) 242, (BLP2)242, and (BLP3) 243, respectively. Top bitlines refer to bitlinesrunning on top of the whole array and running the length of theMFLSUBARY 101. Segment bitlines refer to bitlines running locally withina basic array unit ARYSEG0 290. A set of transistors 230, 231, 232, 233,234, 235, 236, 237 couples respectively segment bitlines (SBL0) 240A and(SBL1) 240B, (SBL2) 241A and (SBL3) 241B, (SBL4) 242A and (SBL5) 242B,(SBL6) 243A and (SBL7) 243B to an inhibit line (VINHSEGO) 274. A line(CL0) 264 is the common line coupled to common lines of the first fourrows of memory cells. A line (CL3) 269 couples to common lines of thelast four rows of memory cells. A set of control gates (CG0) 262, (CG1)263, (CG2) 265, (CG3) 266 couples to control gates of memory cells ofthe first four rows respectively. A set of control gates (CG12) 267,(CG13) 268, (CG14) 270, (CG15) 271 couples to control gates of memorycells of the last four rows, respectively. A pair of inhibit selectlines INHBLB0 272 and INHBLB1 273 couples to gates of transistors 231,233, 235, 237 and transistors 230, 232, 234, 236 respectively. A pair ofbitline select lines (ENBLB0) 260 and (ENBLA0) 261 couples to gates oftransistors 221, 223, 225, 227 and transistors 220, 222, 224, 226,respectively.

Multiple units of the basic array unit (ARYSEG0) 290 are tiled togetherto make up one sub-array (MFLSUBARY) 101 as shown in FIG. 3C. Andmultiples of such (MFLSUBARY) 101 are tiled horizontally to make up thefinal 8192 columns for a total of 32768×8192=268,435,460 physical memorycells, or called 256 mega cells. The logical array size is 256 megacells×4 bits per cell=1 giga bits if 4-bit digital multilevel memorycell is used or 256 mega cells×8 bits per cell=2 giga bits if 8-bitdigital multilevel memory cell is used. The top bitlines (BLP0) 240,(BLP1) 241, (BLP2) 242, and (BLP3) 243 run from the top of the array tothe bottom of the array. The segment bitlines (SBL0) 240A, (SBL1) 240B,(SBL2) 241A, (SBL3) 241B, (SBL4) 242A, (SBL5) 242B, (SBL6) 243A, and(SBL7) 243B only run as long as the number of rows within a segment, forexample, 128 rows. Hence the capacitance contributed from each segmentbitline is very small, e.g., 0.15 pF.

The layout arrangement of the top bitlines 240–243 in relative positionwith each other and with respect to the segment bitlines (SBL0) 240A,(SBL1) 240B, (SBL2) 241A, (SBL3) 241B, (SBL4) 242A, (SBL5) 242B, (SBL6)243A, (SBL7) 243B are especially advantageous in reducing the bitlinecapacitance. The purpose is to make the top bitlines as truly floatingas possible, hence the name of truly-floating-bitline scheme.

In an embodiment as shown in FIG. 5A, lines 240, 241, and 242 are in themiddle, sandwiched between lines 240A, 240B, 241A and 241B in the bottomand lines (CL0) 264 in the top. Furthermore, line 240 is on top of thespacing between lines 240A and 240B and line 241 is on top of thespacing between lines 241A and 241B. This has the benefit of reducingsignificantly the bottom plane capacitance of line 240 and line 241since the oxide below each line is almost doubled. The lines 240 and 241could be positioned on top of lines 240A and 241A, respectively, whenthe sidewall capacitance reduction outweighs the benefit of the bottomplane capacitance reduction. The sidewall capacitance refers to thecapacitance resulting from the vertical walls of a line, the bottomplane capacitance refers to the capacitance from the bottom of a line,and the top plane capacitance refers to the capacitance from the top ofa line.

In another embodiment, as shown in FIG. 5B, the top bitlines 240–242have been positioned all the way to the top metal of a multi-layer metalintegrated circuit system. For example, for a 5-layer metal integratedcircuit system, the top bitlines are metal 5 layer. This avoids the topplane capacitance of the top bitlines 240–242. This also reduces thebottom plane capacitance of the top bitlines 240–242 by a factor of asmuch as 4 if metal 5 is used. The reduction factor of 4 is due to theoxide below the line increasing by a factor of about as much as 4. Alsosince the top bitlines 240–242 are spaced further apart as compared tothe segment bitlines, the sidewall capacitance is reduced significantly.The top bitlines are now almost floating on top of the array. The endeffect is more than on order of magnitude reduction in bitlinecapacitance. Also since the top bitlines 240–242 spacing are relaxed,the width of the top metal lines can be made larger to reduce the metalbitline resistance.

The reduction in bitline capacitance results in a corresponding increasein speed. To help increase the speed in programming, abitline-stabilization-assisted operating method can be applied and isdescribed as follows. At the beginning of the programming cycle, abitline stabilization control signal is used to set all the bitlines toa predetermined voltage VBLPRE, e.g., 0.4–0.8 V. Then high voltage VCLis applied to selected memory common lines for programming. Now thebitlines only have to move partially to a final voltage. This speeds upthe TBL timing.

There is an important transient effect related to bitline capacitance inprogramming. For high speed writing, each program cycle takes time inthe microsecond range. The program bias condition for a memory cell iscontrol gate voltage VCGP,=˜0.7–2.5 V, bitline cell currentIpcell,=˜50–500 nA, and common line voltage VCL going from a low,=˜0 V,to a high programming voltage,=˜8–13 V. As the VCL ramps from a low to ahigh voltage, there is a transient current flowing through the memorycell to charge up the bitline node capacitance. This transient currentflowing through the cell contributes to the cell programming in additionto the programming current Ipcell. Prior art CHE programming would notbe bothered with this effect since the additional transient programmingcurrent is small compared to the actual programming current. However,for a very fine programming voltage level control as required for highbits per cell, this effect will cause the programming level to beuncontrollable, making the multilevel memory system useless. Thefollowing example is given to appreciate the magnitude of this transientcurrent. Assuming program VCL ramp time=1 μs, CBL=1 pF, the voltage thebitline has to slew=1 V, then, by equation (2), I=CV/T=1 pF×1 V/1 μs=1μA, which can be 10× the programming current. Hence a method is neededto reduce the transient programming current.

Two approaches are shown in FIG. 5C to reduce this transient phenomenon.In one embodiment, 2-step ramp rate control approach greatly reducesthis transient effect without prolonging the programming time asfollows. First VCL ramps fast during TRP1 to an intermediate voltageVCLINT, e.g., 2–6 V, then VCL stays at an intermediate voltage for afinite time TVCLINT, then VCL ramps slow during TRP2 to a final voltageVCLFIN. The first fast ramp with the flat intermediate time TVCLINT willlet transient current flowing through the cell to stabilize most of thecell capacitances such as CBL in a short time and at sufficiently lowVCL voltage so that insignificant programming takes place while thetransient current is flowing. The TRP1 is made fast to consume littleprogramming time. The second slow ramp then brings the cell to a finalprogramming voltage without affecting the programming rate since verylittle current is flowing through the cell in the second ramp.

Another embodiment of the ramp rate control is a fast-slow ramp ratecontrol approach. VCL first ramps fast during TRP1 to an intermediatevoltage VCLINT, then VCL ramps slow during TRP2 to a final voltageVCLFIN. The first ramp TRP1 is faster than that of the TRP2 ramp toallow the transient current during the first ramp TRP1 to stabilizequickly all the cell capacitances while VCL is low enough to not causesignificant programming.

The ramp rate can be generated by a RC network, meaning the rate iscontrolled by a certain capacitance multiplied by a certain resistance,or by a CV/I network, meaning the rate is controlled by a certaincapacitance multiplied by a voltage swing divided by a certain biascurrent. Further, the ramp rate can be programmable by programmablefuses as a function of bitline capacitance to optimize the programmingtime without introducing adverse transient current. That is the ramprate is made to be faster for smaller bitline capacitance.

The common line CL0 264 is common to four rows of memory cells for thefollowing reason. Allowing 4 mV voltage drop along the CL line duringprogramming to avoid programming error as described previously, with1024 cells operating simultaneously with 0.1 μa drawn per cell, thevoltage drop by equation (1) is, dVCLP=4 mV=0.5*(1024) (1025) R8cell*0.1μa, hence R8cell=76 milliohms. For a typical CL line with the line widthhalf as wide as the memory cell, the CL resistance per cell is=˜80milliohms, for 8 cells in series, R8cell is 8×80=640 milliohms, which ismuch greater than 76 milliohms. Hence by making CL line 264 four memorycells wide, R8cells is=˜80 milliohms. The reason the width of the lineCL 264 cannot be made arbitrarily large is due to the program disturb.As the high voltage is applied to CL line 264 in programming, all thecells connected to the CL line 264 will see the VCL voltage whether theyare selected for programming or not. The more cells connected to thesame CL line, the longer time for the disturb for the unselected cells.

Shown in FIG. 4A are the metal strapping lines (CL0STRAP) 264S and (CL3STRAP) 269S of the common lines that connect the diffusion common linesto the metal common lines. The metal strapping could be done every 8,16, or 32 memory cells depending on an allowable voltage drop along thecommon line diffusion inside the strapping. This voltage drop depends onthe diffusion common line resistance for a given operating current.

An alternative method that mitigates the voltage drop problem along thecommon line in the program cycle is by theconstant-total-current-program scheme. Namely by keeping the same totalcurrent flowing all the time independent of whether the cells have beenverified or not, the common line voltage drop is kept constant duringprogramming. This could be done for example, by adding additionalswitching transistors in the array every 8, 16, 32, or 64 memory cellsand switching into the CL line the current equivalent to the currentfrom verified cells.

Table 1 shows the operating conditions for the memory array in read,erase, and program. The array operating conditions are shown for thecell 200 of the block ARY1BLK 290A in FIG. 4A, of a selected page forread and program. The selected cell 200 is one cell out of 1024 selectedcells within a selected page. The other 1023 selected cells belong tothe other 1023 ARYSEG0 290 connected horizontally. The array operatingconditions are also shown for all cells connected to CL0 264 for erase.

As shown in Table 1, the operating conditions are such that all theunselected memory cells see no voltage other than 0 volts. This reducessignificantly the power consumption. This is also particularlyadvantageous for improved speed in very high-density memory chips sinceall the necessary driver circuits only see the loading from the selectedmemory cells. The loading from the whole array is tremendous due tolarge number of transistors in array, e.g., 256 million transistors,with its tremendous diffusion, metal and poly interconnect parasitics.For example, one bitline capacitance, CBL is 25 pF, with 8192 bitlinesthe total bitline capacitance is 8192×25 pF=204 nF. This would require atremendous amount of power during signal switching, for example, toinhibit all the bitlines during programming. Also not shown in Table 1,the unselected control signals ENBLAs, ENBLBs, INHBLAs, and INHBLBs forunselected array units ARYSEG0 290 only see 0 or VDD but not themultiplied high voltage. This again saves significant power andincreases speed due to no loading from unselected control circuits.

Another factor that is reduced greatly is the excessive leakage currentfrom the bitline to ground due to junction leakage, bitline to bitlineleakage, band-to-band tunneling, and cell subthreshold conduction. Forexample, for a typical leakage of 10 pA per cell, with 16,384 cells perbitline, the total leakage is 164 nA, which is greater than Ipcell=100nA. This implies that the multilevel programming will be uncontrolleddue to the uncontrollable excessive leakage current contributing to thecontrolled programming current Ipcell. With the inhibit and segmentationscheme, the total leakage current is reduced to 128×10 pA=1.28 nA, whichis much less than Ipcell=100 nA.

FIG. 4B shows an alternative array architecture in which the decodedinhibit line VINHSEGO1 274B is shared between any two adjacent segments.This has the benefit of reducing the number of inhibit lines in thearray.

FIG. 4C shows an alternative array architecture in which the inhibitline VINH 999 is shared for all the segments. This has the benefit ofsharing one inhibit line for the whole array.

FIG. 4D shows an alternative array architecture in which a set ofinhibit select line INHBLA1-3 and INHBLB1-3 275 to 280 are used toinhibit all segment bitlines except the selected segment bitline. VINH999 is shared for all the segments. The operating method makes use of asegment cascading scheme that is described as follows. To even isolatethe bitline capacitance further, bitline select transistors 220–227 arealso used as cascading transistors in programming in addition to theselect and inhibit function. In programming, cell 200 for example, thevoltage on line 261 is initially pulsed high to pass inhibit voltageVINH 999 from a page select (PSELS) 120S into the selected segmentbitline (SBL0) 240A. Then the voltage on line ENBLA0 261 is pulsed to acascading voltage (VPBCAS), e.g., 1 V. A precharge signal then chargesthe selected top bitline (BLP0) 240 to 0.3V. The final voltage on thetop bitline (BLP0) 240 is=˜0.3 V since 1V−VT=˜0.3 V. Hence the voltageon line BLP0 240 no longer changes during programming. The voltage onthe segment bitline, however, still changes as VCL is applied andstabilized. But the capacitance on the segment bitline is minimal,=˜0.15pF. Here the operating method just described could also apply to thearray shown in FIG. 4A but the inhibit voltages on the unselectedsegment bitlines are floating. The array shown in FIG. 4D just makessure all the unselected segment bitlines are kept at a constant inhibitvoltage (VINH) 999.

FIG. 4E shows another array suitable for the method just describedabove. It needs a set of 4 additional lines (INHBLAB0-3) 281–284 and aset of 8 additional transistors 240I–247I for inhibit decoding. Howeveradditional transistors 240I–247I occupy less die area than that requiredfor additional inhibit decoding lines 275–280 in FIG. 4D.

FIG. 4F shows an array architecture similar to that in FIG. 4A with theinhibit transistors physically at the top of the segment array.

Note that it is possible to do one top bitline per one segmented bitlinein the ARYSEG0 290. In this case, the sidewall capacitance from one topbitline to adjacent top bitlines increases due to reduced spacingbetween the top bitline and the adjacent top bitlines.

Note that it is also possible to do one top bitline per more than twosegmented bitlines in the ARYSEG0 290. In this case, more decodingtransistors are needed in the array to select one segmented bitline outof more than two segmented bitlines, which leads to more die size.However the sidewall capacitance from one top bitline to adjacent topbitlines decreases due to increased spacing between the top bitline andthe adjacent top bitlines. This reduction of capacitance may not besignificant if the spacing is already wide enough.

An alternative embodiment of reducing the bitline capacitance is byhierarchical interconnect segmentation that is an extension over theprevious concept as follows. A first segment bitline running in firstlayer of metal couples to a plurality of memory cells. A second segmentbitline running in second layer of metal is coupled to a plurality offirst segment bitlines by bitline segment transistors through viasbetween metal 1 and metal 2. Third segment bitline running in thirdlayer of metal is coupled to a plurality of second segment bitlines byother bitline segment transistors through vias between metal 1 and metal2 and metal 3. This can continue to higher metal layers. This approachallows optimization of horizontal spacing, vertical spacing,interconnect width, and interconnect length between different layers ofinterconnect metals for minimum capacitive coupling between metalinterconnect lines. This results in further reduced bitline capacitance.

TABLE 1 Array Operating Conditions READ ERASE PROGRAM SELECTED SEGMENTS:CG0 3–6 V 8–13 V 0.7–2.5 V CG1, 2, 3 0 8–13 V CG4-15 0 0 0 Rest of all 00 0 CG lines CL0 2–3 V 0 4–13 V CL1, 2, 3 0 0 0 Rest of all 0 0 0 CLlines BL0, 8, 16 . . . 0 TO 2–3 V FL or 0 V 0–0.8 V BL1–7, 9–15, VINHVINH VINH 17–23, . . . UNSELECTED SEGMENTS: All CG lines 0 V 0 V 0 V AllCL lines 0 V 0 V 0 V All BL lines 0 V 0 V 0 VMultilevel Memory Decoding

FIG. 6 shows the block diagram of the multilevel decoding scheme. Theinvention provides precision voltages with millivolt control tolerancesto the memory array over temperature, process corners, and power supplyvariation. The invention provides these voltages in an efficient manner,meaning deliver power where it is needed and reducing the output loadingthrough circuit configuration. The invention also provides a multilevelprecision decoding circuit with minimum area overhead.

As discussed in the array architecture section, the voltage drop alongthe common line would cause a programming error as well as sense errorin read. Hence the drop is brought down to a manageable level. Bypartitioning a common line into small line sections, with drivers onboth sides of each of the line sections, the voltage drop is reduced.However, prior art partition would cause a tremendous area penalty dueto the large amount of decoding lines and circuits. This inventionprovides an enhanced decoding circuit by routing the interconnect in thehigher metal layers and by using circuit configurations suitable formultilevel decoding.

The block (VCGCLPRED) 156 has been expanded to include sub-blocksinside. Common line predecoder and driver (XCLPREDRV) 950 providepredecoded common lines with precision voltages to regular memory commonlines in block 130 and 132. A common line predecoder and driver(XCLSPREDRV) 954 provides predecoded common lines with precisionvoltages to spare memory common lines in block 134. The circuit block954 is functional equivalent to circuit 950. A control gate predecoder(XCGPREDEC) 951 provides predecoded control gate lines to block 130. Aspare control gate predecoder (XCGSPREDEC) 952 provides predecodedcontrol gate lines to block 134. A bitline predecoder (BLXDEC) 953provides predecoded bitlines to block (MLMDEC) 130. All other circuitblocks have been described in association with FIG. 2A.

FIG. 7 shows one segmented decoder (RD1SEG) 300. The RD1SEG 300 selectsor deselects a plurality of basic array unit (ARYSEG0) 290 connectedhorizontally. The RD1SEG 300 includes a circuit segmented supply decoder(RDSGPSDEC) 301, a segmented bitline decoder (RDSGBLDEC) 302, asegmented common line pre-decoder (RDSGCLPDEC) 302B, a segmented inhibitdecoder (RDSGINHDEC) 303, and multiples of a sub-block decoder(RD1SUBBLK) 304. The RDSGPSDEC 301 decodes the high voltage supply foreach segmented decoder (RDLSEG 300). The high voltage supplies for theunselected segmented decoders (RD1SEG) 300 are disabled and hence poweris minimized due to much less loading and die size is reduced due to asmaller voltage multiplier. The RDSGBLDEC 302 couples the segmentbitlines to the top bitlines when selected. The RDSGINHDEC 303 couplesthe inhibit voltage (VINH) 999 to the appropriate bitlines of theselected array units (ARYSEG) 290 when selected or unselected asdescribed later in FIG. 9B. The RD1SUBBLK 304 enables appropriatecontrol gates and common lines for the memory cells.

FIG. 8 shows details of the power supply decoder (RDSGPSDEC) 301. Line(NI) 310 and (OI) 311 are predecoded address lines coming from theaddress predecoder block (XPREDEC) 154. Line ENVSUPDEC 312 is a globalenable signal for disabling or enabling all the supply decoders. A NANDgate 315 is a typical 3-input NAND gate with an output line (ENB) 313.An inverter 316 is a typical inverter with input line (ENB) 313 and anoutput line 314. A high voltage level shifter (HVLS1) 317 shifts logicsignal EN 314 into high voltage complementary output signal lines(ENVSUPB) 318 and (ENVSUP) 319. A line (VXRGND) 333 is a low voltageline for (HVLS1) 317. A line (VHSUPPLY) 777 is a precisely regulatedhigh voltage supply for the decoding. A line (VMSUPPLY) 666 is anotherprecisely regulated high voltage supply. A transistor PMOS 322 couplesthe high voltage supply (VHSUPPLY) 777 into line (VHSUPPLYSG) 328 whenthe RDSGPSDEC 301 is selected. Transistors PMOS 323 and 324 coupleregular voltage supply (VDD) 1111 into line (VHSUPPLYSG) 328 when theRDSGPSDEC 301 is deselected. A transistor PMOS 325 couples another highvoltage supply (VMSUPPLY) 666 into line (VMSUPPLYSG) 329 when theRDSGPSDEC 301 is selected. The voltage level on line (VMSUPPLY) 666,e.g., 5–10V, is such that in read the bitline select transistors in thememory array are heavily overdriven to reduce their on resistance, whichresults in insignificant sense error. Transistors PMOS 326 and 327couple regular voltage supply (VDD) 1111 into line (VMSUPPLYSG) 329 whenthe RDSGPSDEC 301 is deselected. The PMOS 323 and 326 have their wellsconnected to line (VDD) 1111. The PMOS 324 and 327 have their wellsconnected to the VHSUPPLYSG 328 and VMSUPPLYSG 329, respectively. Theconnection of their wells is done to avoid source and drain junctiondiodes turning on during the switching.

FIG. 9A shows details of the segmented bitline select decoder(RDSGBLDEC) 302. Line (ENVSUP) 319 and line (ENBLAVH) 341 connected tothe gates of transistors 360 and 361, respectively, are used to couplevoltage on line VMSUPPLYSG 329 into line ENBLA 369. Either transistor362 with line (ENB) 313 on its gate or transistor 363 with line(ENBLBVL) 342 on its gate is used to couple line (ENBLA) 369 to line(VXRGND) 333. Similarly transistors 364 and 365 together with lines(ENVSUP) 319 and line (ENBLBVH) 343, respectively, on their gates areused to couple voltage on line (VMSUPPLYSG) 329 into line (ENBLB) 371.Either transistor 366 with line (ENB) 313 on its gate or transistor 367with line ENBLAVL 340 on its gate are used to couple line (ENBLB) 371 toline (VXRGND) 333. The voltage level on line (VHSUPPLY) 777 in the block(RDSGPSDEC) 301, e.g., 7–12 V, is such that the transistors 360, 361,364, 365 couple, with minimal loss, the voltage from VMSUPPLYSG 329 intolines (ENBLA) 369 and (ENBLB) 371. The deselect transistors 362, 363,366, and 367 have their gates coupled only to the low voltage signalsinstead of the high voltage control signals as conventionally done. Thiscircuit configuration has the benefit of reducing significantly theloading for the high voltage supply (VHSUPPLY) 777. This circuitconfiguration is applied throughout all the decoding circuits.

FIG. 9B shows details of the segmented inhibit select decoder(RDSGINHDEC) 303. Either transistor 350 with line (ENVSUPB) 318 on itsgate or transistor 353 with line (ENBLBVH) 343 on its gate couples thevoltage on line (VMSUPPLYSG) 329 to line (INHBLA) 345. Transistors 351and 352 together with lines (EN) 314 and (ENBLAVL) 340, respectively, ontheir gates are used to couple line (INHBLA) 345 to line (VXRGND) 333.Similarly either transistor 354 with line (ENVSUPB) 318 on its gate ortransistor 357 with line (ENBLAVH) 341 on its gate is used to couple thevoltage on line (VMSUPPLYSG) 329 to line (INHBLB) 347. Transistors 355and 356 together with lines (EN) 314 and line (ENBLBVL) 342 respectivelyon their gates are used to couple line (INHBLB) 347 to line (VXRGND)333. Transistor 358 with line (ENVSUP) 319 on its gate is used to couplethe inhibit voltage on line (VINH) 999 to line (VINHSEG) 349. Transistor359 with line (ENB) 313 on its gate is used to couple the voltage online (VINHSEG) 349 to line (VXRGND) 333. Similar to the circuitconfiguration in the RDSGBLDEC 302, the low voltage signals are used forsignal deselection.

The circuit blocks RDSGPSDEC 301, RDSGBLDEC 302, RDSGINHDEC 303, andRD1SUBBLK 304 are used in the array as shown in FIG. 4A for arrayselection and inhibit decoding.

FIG. 9C shows a predecoded common line segmented decoder (RDSGCLPDEC)302B for lines (CLP0-3) 445A–D. Lines (CLP0-3) 445A–D come from a commonline pre-decoder (XCLPREDRV) 950. The purpose of this circuit(RDSGCLPDEC) 302B is to greatly reduce the capacitive loading on linesCLP0-3 seen by the common line pre-decoder (XCLPREDRV) 950. Lines(CLPS0-3) 456A–D are the output lines. Transistors 438A–D with line(ENVSUP) 319 on their gates are used to couple lines (CLP0-3) 445A–D tolines (CLPS0-3) 456A–D, respectively. Transistors 439A–D with line (ENB)313 on their gates are used to couple lines (CLPS0-3) 456A–D to line(VXCLGND) 5555. This concept of segmented loading could also be appliedto predecoded control gates CGP0-15.

FIG. 10 shows details of the sub-block decoder (RD1SUBLK) 304, thatincludes a circuit block 304A and a circuit block 304B. The bloc6tgk304A includes a NAND gate 412, an inverter 413, and a high voltage levelshifter (HVLSX) 418. The 3-input NAND gate 412 is used for addressdecoding. Line (ENB4) 414 is its output. Lines (MI) 410, (NI) 310, and(OI) 311 are predecoded address lines coming from the addresspre-decoder (XPREDEC) 154. The inverter 413 inverts line (ENB4) 414 intoline (EN4) 415. The high voltage level shift (HVLSX) 418 is used toshift the logic signal EN4 415 into the high voltage output signal(ENHV4BLK) 417. Line (VHSUP) 770 supplies high voltage for the levelshifter (HVLSX) 418. Line (VHSUP) 770 couples to line (VHSUPLYSG) 328 ofcircuit block (RDSGPSDEC) 301. The circuit block 304B including a set offour circuit blocks (RD4CG1CL) 416 provides control signals for controlgates (CG) and common lines (CL). Lines CG[0:15] 422A–P couple to 16rows of memory cells, for example, lines 262, 263, 265–268, 270, 271 ofthe block (ARY1BLK) 290A in FIG. 4A. Lines CL[0:3] 423A–D couple to 4shared common lines of memory cells, for example, lines 264 and 269 ofthe block ARY1BLK 290A in FIG. 4A. Lines CGP[0:15] 420A–P are predecodedcontrol gate lines coming from the control gate pre-decoder (XCGPREDEC)951. Lines CLPS[0:3] 456A–D are predecoded common lines coming fromblock RDSGCLPDEC 302B. Line (VXCGGND) 444 is a line for control gate(CG) deselection. Line (VXCLGND) 5555 is a line for common line (CL)deselection.

FIG. 11A shows details of circuit block (RD4CG1CL) 416. Transistors 430,432, 434, 436 together with lines (CGP0) 440, line (CGP1) 441, line(CGP2) 442, line (CGP3) 443, respectively, on their drains are used tocouple these lines 440–443 to output line (CG0) 450, line (CG1) 451,line (CG2) 452, and line (CG3) 453, respectively. Lines (CGP0-CGP3)440–443 come from a control gate predecoder (XCGPREDEC) 951. Transistor438 is used to couple line (CLPS0) 456A to line (CL0) 454. Transistor439 is used to couple line (CL0) 454 to line (VXCLGND) 5555. Line(ENHVLBLK) 446 couples high voltage into the gates of transistors 430,432, 434, and 436. Line (ENB1BLK) 447 couples lines (CG0-3) 450–453 tothe line (VXCGGND) 444 through transistors 431, 433, 435, and 437,respectively, and couples line (CL0) 454 to line (VXCLGND) 5555 throughtransistor 439. The lines (ENHV1BLK) 446 and (ENB1BLK) 447 are coupledrespectively to lines (ENHV4BLK) 417 and (ENB4) 414 generated by circuitblock 304.

Four common lines of memory cells are coupled together to one decodedcommon line CL as shown in the block (ARYSEG0) 290 in FIG. 4A. Fourblocks of the RD4CG1CL 416 are used to provide array block selection asshown in the block (ARYSEG0) 290 in FIG. 10. One array block is definedas including 16 rows and 4 common lines of memory cells. One array blockincludes a plurality of blocks (ARY1BLK) 290A connected horizontally.

The lines (VXRGND) 333, (VXCLGND) 5555, and (VXCGGND) 444 could beindividually controlled to be biased at different voltage levels duringerase, read, and program to optimize circuit functionality, forinstance, to increase the breakdown or to reduce the leakage of MOSdecoding transistors.

Note that the same transistors are used for decoding in erase, read, andprogram operation. In conventional decoding, read decoding is isolatedfrom erase and program decoding since read decoding requires only lowvoltage and hence the decoding size can be optimized for read speed.Here all decoding is combined together to minimize the die size. Furtherall decoding is done by NMOS transistors instead of by both PMOS andNMOS transistors as conventionally done. This has the benefit ofreducing the capacitive loading. This is so because in deselection onePMOS presents itself as a gate capacitor load while one NMOS onlypresents itself as a source or drain overlap capacitor load, which ismuch smaller than a gate capacitor load. Low capacitive loading leads toless power consumption for NMOS decoding. This is against conventionalwisdom, which holds that a CMOS circuit is more power efficient than aNMOS circuit.

FIG. 11B shows an alternative circuit block (RD4CG1CL) 416 with adiode-connected transistor 438F. The transistor 438F provides feedbacksignal (CLK) 445F for a Kelvin type connection to a circuit driverinside the block (XCLPREDRV) 950. A Kelvin connection line consumesminimal (or no) DC current. A Kelvin connection allows a circuit driversuch as a common line circuit driver to stabilize its output signal at adesired voltage level based on feedback voltage from the Kelvinconnection line. This Kelvin connection line (CLK) 445F is connected toother Kelvin connection lines vertically. This is possible since onlyone common line is on at any given time. Once a common line is selected,this common line will take control of the CLK 445F line since thediode-connected transistor will be forward biased and otherdiode-connected transistors on the rest of the common lines will bereverse biased. This will be known as winner-take-all Kelvin decoder.This winner-take-all Kelvin decoder will ensure a predetermined voltageon the line (CL0) 454 will be stable all the time over varying load,process corners, temperature, and power supply variation with minimumcost. The stable voltage on the common line is required to not introducesignificant voltage error in program or in read as described previouslyin the description of the multilevel array architecture.

FIG. 11C shows a circuit block (RD1CL) 304C, which is used in a commonline segmentation scheme with the array partitioning shown in FIG. 12 toreduce the voltage drop along the common lines. In an embodiment, onecommon line (CL) is connected together across the full array with aplurality of blocks (RD1CL) 304C driving the same common line (CL).Transistor 438S with line (ENHV1BLK) 446 on its gate couples line(CLPS0S) 456AS to line (CL0) 454. Line (CL0) 454 of this circuit block304C is the same line (CL0) 454 of the circuit block (RD4CG1CL) 416. Adeselect transistor 439S with line (ENB1BLK) 447 couples line (CL0) 454to line (VXCLGND) 5555. The transistor 439S is optional in this circuitsince the function of coupling line (CL0) 454 to line (VXCLGND) 5555 isalready provided by the transistor 439 in the RD4CG1CL 416. Thetransistor 439S provides additional drive ability in addition to that ofthe transistor 439. Line (CLPS0S) 456AS couples to a common linepre-decoder (XCLPREDRV) 950. The winner-take-all Kelvin decoding canalso be used here. The control signals (ENHV4BLK) 417 and (ENB4) 414shown in the block (RD1SUBBLK) 304 couple to control signals (ENHVLBLK)446 and (ENB1BLK) 447, respectively. The control signals (ENHV4BLK) 417and (ENB4) 414 are fed through the memory array as shown in FIG. 12. Inan alternate embodiment, one common line is divided into many separatecommon lines across the full array. These separate common lines are notconnected to each other. In this case, each separate common line isdriven on both sides by two blocks (RD1CL) 304C or by a (RD1CL) 304C anda (RD4CG1CL) 416. Common line segmentation is described more in detailbelow in description associated with FIG. 12.

FIG. 12 shows a feedthrough-to-memory and feedthrough-to-driver schemetogether with the common line segmentation to deliver precise voltagesfor memory cells as described in the following. The feedthrough schemeexploits the multi-layer metal interconnect to reduce the circuitcomplexity and die size and to enable innovative circuit configurations.A conventional flash memory system typically only uses up to a maximumof 2 metal layers and hence is limited in core interconnect schemepossibilities. This feedthrough scheme is made possible by three or moremetal layers.

The block (MLMDECS) 132, shown in FIG. 12 and also in FIG. 3A, includesa plurality of the blocks (RDSGCLPDEC) 302B and a plurality of theblocks (RD1CL) 304C. Only one block (RDSGCLPDEC) 302B and one block(RD1CL) 304C per block 132 are shown in FIG. 12 for clarity. Otherblocks have similar connections. The block (MLMDEC) 130, shown in FIG.12 and also in FIG. 3A, includes a plurality of the blocks (RD1SEG) 300.The block RD1SEG 300 includes a block (RDSGPSDEC) 301 and a plurality ofblocks (RD1SUBBLK) 304. Only the block (RDSGPSDEC) 301 and one block(RD1SUBBLK) 304 inside one block RDLSEG 300 are shown in FIG. 12 forclarity. Other blocks have similar connections.

The feedthrough-to-memory uses a single driver to drive both left andright sides of a memory array. The layout of row decoding circuits suchas of the block (RD1SUBBLK) 304 is very dense because of the limitedheight of a typical advanced memory cell, e.g., 0.5–1 μm per cellheight, and the very wide width of each decoding transistor, e.g., 20–50μm, due to their required precision multilevel drive ability. This makesit extremely difficult to route the required lines from the right sideacross the active circuit of this row decoding circuit to the left sidewith limited layers of metal interconnect. As shown in FIG. 10, thecontrol lines CG[0:15] 422A–P and common lines CL [0:3] 423A–D providesthe control signals to the memory cells on the right side as well as thememory cells on the left side. This is also shown in FIG. 12 in block304B with lines pointing to the right as well as to the left. Similarlyit also shows the control lines from circuit block 304A and 304C drivingboth sides. The feedthrough-to-memory scheme also shows predecoded highvoltage lines (ENHV4BLK) 417 and (ENVSUP) 319 and predecoded low voltagelines (ENB) 313 and (ENB4) 414 being fed through the memory by runningon top of the memory, for example, in metal 4, without interfering withthe memory cells underneath. Other control lines could also be fedthrough the memory. Again this is achievable by three or more metallayers which allow a different circuit configuration with minimal activearea. The circuit block 304C is the precision voltage driver for thecommon lines CL of the memory cells in addition to the circuit block304B. The feedthrough-to-driver scheme shows control signals fromcircuit blocks 304B and 304A being fed through the memory array to theprecision voltage drivers 304C.

The common line segmentation is also shown in FIG. 12. Each metal commonline runs the length of the memory core horizontally across the fullarray with seven circuit blocks (RD1CL) 304C and two circuit blocks(RD1SUBBLK) 304 driving the same common line. The voltage drop acrossone common line is thus divided into eight voltage drop segments. Eachvoltage drop segment belongs to each common line of each sub-array block(MFLSUBARY) 101. Within each voltage drop segment, the voltage value onthe left side is same as the voltage value on the right side of thevoltage drop segment and the lowest voltage value is in the middle ofthe voltage drop segment. This is because there is a precision circuitdriver (RDLCL) 304C or (RD4CG1CL) 416 on each side of the voltage dropsegment. One alternative embodiment of the common line segmentationscheme is to have these common lines physically divided into eightseparate common lines. That is, each sub-array block (MFLSUBARY) 101shown in FIG. 12 has its separate common line. However, in this case,the deselect transistor 439S in the block (RD1CL) 304C is no longeroptional but necessary to deselect each separated common line.

The voltage level on the control gates is controlled by the voltage onthe lines (CGP[0:15]) 420A–P in circuit block 304. The voltage on lines(CGP[0:15]) 420A–P are in turn controlled by a precise bandgap-referredregulated voltage. Hence precision voltage level is provided at thememory control gates. The voltage level on the common lines iscontrolled by the voltage on the predecoded common lines (CLP[0:3])421A–D in circuit block 304. The voltage on lines (CLP[0:3]) 421A–D arein turn controlled by a precise bandgap-referred regulated voltage foreach common line driver. Hence precision voltage level is provided atthe memory common lines. The programming and sensing current bias arealso bandgap-referred; hence they are highly stable.

Note that in FIG. 12 an alternative embodiment is to share one block(RDSGPSDEC) 301 or 304A across the full array by doing feedthrough ofthe outputs of (RDSGPSDEC) 301 or 304A across the full memory array. Inthis case the drive ability of the driver circuit inside block(RDSGPSDEC) 301 or 304A should be adequately designed to drive the longinterconnect lines.

Note that in FIG. 10 an alternative embodiment is to have a separateblock (RD4CG1CL) 416 for driving the right side of an array and anotherseparate block (RD4CG1CL) 416 for driving the left side of an array.Another alternative embodiment is to share just one CL driver for bothleft and right sides but to have separate control gate CG drivers forthe left side and the right side.

Multilevel Reference System

FIG. 13 shows a block diagram for a multilevel digital memory referencesystem. All the relevant blocks have been described in association withprevious figures. The highlighted blocks 106, 116, 126, and 146 with thehighlighted lines (VREF0-15) 760–775 are shown to show the referencesystem in relation to the physical position of the array and y-drivers.The physical position of the reference array corresponding to variousschemes is explained in the following description.

FIG. 14 shows details of a multilevel digital memory reference system. Areference circuit block (VREFGEN) 719 is used to provide all referencevoltage levels for erasing, programming, sensing, margin tests, andproduction tests. Shown are reference levels for reference cells(VREFR0-15) 700–715 and reference levels for data cells (VREFD0-15)720–735. Data cells refer to memory cells that store digital data. A 16level multilevel flash cell is assumed for this discussion. A flashreference array (MFLASHREF) 106 includes a plurality of blocks(MFLASHREFS) 106A. A block (MFLASHREFS) 106A includes a plurality ofreference memory cells. A reference page select 126A is used to selectthe reference cells in the blocks (MFLASHREFS) 106A associated with aselected page. Each block 126A selects one reference cell in onecorresponding block (MFLASHREFS) 106A. For each selected page, there are16 blocks 126A selecting 16 reference cells in 16 corresponding blocks(MFLASHREFS) 106A. The 16 selected reference cells makes up one pagereference.

A buffer (VRBUFFER) 750 and a comparator 801 are inside a block(REFYDRVS) 116S. The buffer (VRBUFFER) 750 is used to drive eachreference level of (VREF0-15) 760–775 for all the y-drivers. A buffercircuit without offset auto zero 750A is used to isolate the referencecell from all capacitance from auxiliary circuits. The offset auto zerocancels out the voltage offset of an analog buffer. The voltage offsetof an analog buffer is typically uncontrollable and is caused bythreshold voltage mismatch, transistor transconductance mismatch, andsystematic offset. This voltage offset would cause an uncertainty in thereference voltage, which would degrade the margin of one voltage levelwith respect to another voltage level. Line (VBUFO) 781 is used toverify a reference cell is programmed to one desired reference level outof 16 possible reference levels. Line (VBUFO) 781 is used instead of thedirect memory cell output for verifying in the verify cycle. This is toinclude the buffer offset from buffer 750A in the verifying process. Thecomparator 801 is used to do the actual comparison in verify. A bufferwith offset auto zero 750B is used to drive a reference level. Variousvoltage levels needed for multilevel algorithm are also generated by thebuffer 750B with switch capacitor technique. The auto zero is needed tozero out the offset of this buffer since a typical buffer offset is10–20 mV. This voltage amount if not canceled out would degrade themargin of a reference level, which effectively reduces the voltagemargin for each level. Capacitors are needed to accomplish the auto zeroand level shifting operation in the buffer 750B. However as described inthe array architecture description, any additional capacitance wouldadversely degrade the write and read speed. Hence buffer 750A isinserted between the reference cell and the buffer 750B so that thereference cell only sees one gate capacitance inside a typical buffer asa capacitor load.

Lines (VREF0-15) 760–775 are the final reference lines driving into allthe y-drivers as needed for verify-program cycles and read cycles.Switch S 750D couples line (VREFD) 720 to the input terminal of buffer750B when one selected page programs for the first time. Switch S 750Ccouples line (VBUFO) 781 to input terminal of buffer 750B when the sameselected page programs for the second time or more without an erase inbetween program. The reason is that for first time programming,reference levels for data cells come from a reference generator VREFGEN719 and for subsequent programming reference levels come from thereference cells in MFLASHREFS 106A.

For the memory system described herein, there are 8 pages for each row,4 rows for each block, and 512 bytes per page with a 4-bit digitalmultilevel memory cell. Since any one page is written or read at anytime a complete reference set of 16 levels is reserved for each pageinstead of for each row. This is done to preserve the operatingconditions through the lifetime of a memory system exactly the same forreference cells as regular data cells. This is done for example to makethe reference and data cells have the same voltage readout drift overtime. For each row, there are 8×16=128 reference cells. This has somesmall die size penalty. The reference cells are written at the same timeas the regular data cells.

After the reference cells are written with the first programmingsequence, if subsequent programming cycles are allowed to write otherdata cells in the same page, the previously programmed reference cellsstay in the program inhibit mode. This is accomplished as shown in FIG.15. A comparator 850 is used to compare a reference voltage from abandgap VREF 851, e.g., 1.2 V, versus a readout voltage from a referencememory cell VREFOUT 852, for example, level 0, e.g., 0.5V. If thereference cell has not been written, VREF 851<VREFOUT 852, then line(REFON) 853 would be low. If the reference cell has been written, VREF851>VREFOUT 852, then line (REFON) 853 would be high indicating that thereference cells have been previously written and the reference cells areinhibited in programming.

For subsequent programming cycles after the first programming cycle, thereference voltages for the data cells come from the reference cells andthe reference voltages are shifted appropriately to place the datavoltages in between the adjacent reference voltages.

The voltage drop along the common line poses a particular problem for amultilevel reference system. Reference cells are needed to track thedata cells over temperature, process, or power supply. But astemperature changes, the voltage drop along the common line changes,which causes a sense error. The voltage drop along the line from one endto the other end follows geometrically as described earlier. That isdepending on position along the common line, the cells experiencedifferent amounts of common line voltage changes, which cause differentvoltage readout shifts due to different voltage amounts being coupledinto the cells. This cannot be corrected by a conventional referencesystem.

FIG. 16 shows a positional linear reference system that corrects thiserror. Assuming the voltage drop along a line is linear and assuming anacceptable voltage shift is DVREF/2, by dividing the voltage dropDVTOTAL 859=VBEG 855−VEND 856, into different voltage segments withequal voltage drop DVREF 858 and by positioning the reference cells 857in the middle of a divided array segment (ARYVSUB1-3) 888A–Ccorresponding to a voltage segment, the maximum voltage difference froma reference cell to a data cell in the beginning or at the end of thevoltage segment is=<DVREF/2. Hence reference correction over temperatureis achieved. It is possible to place the reference cells 857 at thebeginning or the end of a divided array segment (ARYVSUB1-3) 888A–C. Inthis case the maximum voltage difference from a reference cell to a datacell is DVREF instead of DVREF/2 as in the case of positioning thereference array in middle of a divided segment array. Another advantageof placing the reference cells in the middle of a divided array segmentis to minimize the electrical variation due to the edge interface fromthe memory array to peripheral circuits.

FIG. 17 shows a positional reference geometric system basing on theconcepts similar to FIG. 16. In this embodiment, the reference cells 857are not symmetrically but geometrically positioned to correct for thegeometric effect of the voltage drop.

In FIGS. 16 and 17, each full array is divided into three sub-arrays(ARYVSUB1-3) 888A–C and (ARYVSUB4-6) 888D–F respectively. It should benoted that the array could be divided into as many sub-arrays as neededto reduce the voltage error. Also shown in FIGS. 16 and 17, eachsub-array of ARYVSUB1-6 888A–F includes its own complete set ofreference cells in the middle. A complete set of reference cellsprovides all the reference levels, e.g., 16 levels for 4-bit digitalmultilevel cell per page, for all the pages. One row of reference cellsincludes 128 reference cells if each row has 8 pages and each referencecell provides one reference level. An alternative embodiment is to havemore than one reference cell per level, e.g., 4–16 cells per level. Thisaverages out the electrical variation of multiple cells.

FIG. 18 shows a geometric compensation reference system. The objectiveis to simulate the voltage drop in the common line into the referencereadout voltage by attaching similar loading currents to the referencereadout voltage. A resistance R 862 in the reference line is madeequivalent to a resistance R 866 in the common line. A reference loadingcurrent (ICELLR) 868R is made the same as that of ICELL 868. Hence thetotal voltage drop in reference DVREFTOTAL 863,=REFB 860-REFE 861, isequal to DVCLTOTAL 867,=VCLB 864-VCLE 865. It is not necessary to attachthe same number of loading reference currents ICELLR 868R to the numberof ICELL 868. It is only necessary to attach the approximate amount ofthe current loading at appropriate positions to minimize the error to anacceptable level.

One alternative embodiment of the reference system is, instead of using16 reference cells for a 4-bit digital multilevel cell, to use 2 or 4 or8 reference cells to generate 16 reference levels with levelinterpolation. That is from reference levels coming from referencecells, the other reference levels are interpolated by using linear orany other interpolation.

Multilevel Algorithm

FIG. 19A shows various voltages generated and used in one embodiment ofthe invention for program verifying, program upper and lower marginverifying, read sensing and restore high or restore low margin verifyingduring read sensing. The read sensing is advantageously performed in thevoltage-mode but other modes of read sensing are also applicable. Allthe voltages are generated by the V&IREF block 172. VREFR(L) is theprogram verify voltage used to verify program level (L) of a referencecell. VREFD(L) is the program verify voltage used to verify programlevel (L) of a data cell. For example, in a 4 bit per cell storageembodiment there are 16 levels used. It is also possible to use 15levels instead of 16 levels since the extreme low or high levels notneed to be constrained to exact low or high levels but can go to groundor power supply respectively. VREFR0 through VREFR15 are program verifyvoltages used for verifying programming of the reference cells. VREFD0through VREFD 15 are program verify voltages used for verifyingprogramming of the data cells. VUM(L) and VLM(L) are upper and lowerprogram margin voltages respectively for level L. Each level L may haveits own VUM(L) and VLM(L) voltage values. VUM(L) and VLM(L) can each beof different value also for each level L. On the other hand, VUM(L) andVLM(L) can be of the same voltage value for all the levels. VUM(L) andVLM(L) voltages are generated by the block V&IREF 172. VRSTH(L) andVRSTL(L) are RESTORE HIGH and RESTORE LOW margin voltages respectivelyfor level L. Each level L may have its own VRSTH(L) and VRSTL(L) voltagevalue. VRSTH(L) and VRSTL(L) can each be of different value also foreach level L. On the other hand, VRSTH(L) and VRSTL(L) can be of thesame voltage value for all the levels. VRSTH(L) and VRSTL(L) voltagesare generated by the V&IREF 172 block. VCELLR(L) is the voltage readback from a reference cell during read sensing. VCELLD(L) is the voltageread back from a data cell during read sensing. The cross-hatchedregions show the distribution of possible read back voltages during readsensing after reference cells or data cells have been programmed to acertain level L, while using VREFR(L) or VREFD(L) as the program verifyvoltage, respectively. The distributions occur because every cell doesnot have the same programming or read sensing characteristics.

Page Programming Cycle

FIG. 20 shows the flow diagram for one embodiment of the pageprogramming cycle. During a page programming cycle a plurality of memorycells are programmed in parallel. However this algorithm is equallyapplicable for single cell programming. As an example, 4 bit per cell isprogrammed in each cell. First the program command is issued and theaddress of the. page to be programmed is loaded. The data count NC isinitialized. The address loading may be performed through a single or aplurality of address cycles. Program data is input during the DATAINstep and is selectively loaded in the internal latches of a YDRVS 110Sor SYDRVS 114S or RYDRV 112S. Block YDRV 110, SYDRV 114, (RYDRV) 112includes a plurality of YDRVS 10S, SYDRVS 114S, RYDRVS 112Srespectively. Block YDRVS 110S will be described in detail later in thedescription associated with FIG. 26. Data gets loaded into the datalatches of the current YDRVS 110S or SYDRVS 114S selected from theADDRCTR 162 and the BYTEDEC 152. The redundancy control block REDCNTRL186 asserts RED_ADD_TRUE true (YES or Y) or false (NO or N) to signifywhether the current YDRVS 110S or SYDRVS 114S is GOOD or BAD. A YDRVS110S or SYDRVS 114S is GOOD if it has not been flagged as one thatcannot be used to load input data on its data latches. A YDRVS 110S orSYDRVS 114S is BAD if it has been flagged as one that cannot be used toload input data on its data latches. GOOD or BAD YDRVSs or SYDRVSs areflagged during manufacturing testing and the flags are internally storedon non-volatile latches. If RED_ADD_TRUE=NO, meaning current YDRVS 110Sor SYDRVS 114S is GOOD, then a data nibble on the IO[0:3] or IO[4:7] busis placed at the input of the data latches of the current YDRVS 110S orSYDRVS 114S. A data byte consists of 8 digital bits and a data nibbleconsists of 4 digital bits. If RED_ADD_TRUE=Y, meaning current YDRVS110S or SYDRVS 114S is BAD, then the data nibble on the IO[0:3] orIO[4:7] bus is placed at the data latches of the selected RYDRVS 112S.Next, if NEXTDATAIN=Y, the data at the input of the data latches of therespective YDRVS 110S, SYDRVS 114S or RDYRVS 112S is latched. IfNEXTDATAIN=N then the flow waits for the program start command PRG.Next, if the data count NC>MAXNC=not true (N), then NC=NC+1 and the flowloops back to DATAIN step to load in the next data byte. If the datacount NC>MAXNC=true (Y), then the flow goes out of the loop and waitsfor the program start command PRG. The data count MAXNC signifies thenumber of data bytes that are simultaneously programmed in a page. Next,if command PRG is received then page programming begins. If command PRGis not received then the flow loops back to check for NEXTDATAIN. Nodata loading is required for blocks (REFYDRVS) 116S because theirlatches are internally set. A block (REFDRV) 116 includes a plurality ofblocks (REFYDRVS) 116S.

FIG. 21 shows the flow diagram after page programming begins. TheProgram flag=Pass is set and the BUSY signal is set. In anotherembodiment a configuration (fuse) bit initialization is executed to loadin data from fuse non-volatile memory cells to volatile latches locatedin the fuse circuit block (FUSECKT) 182 at this step. The programinhibit mode of all cells in the page being programmed are reset toenable programming. Based on the output B[0:3] of the data latches ofeach YDRVS 110S, SYDRVS 114S or RYDRVS 112S a program verify voltageVREFD(L) is set at the input of the comparator in each of the respectiveYDRVS 110S, SYDRVS 114S or RYDRVS 112S. Based on the output B[0:3] ofthe data latches of each REFYDRVS 116S a program verify voltage VREFR(L)is set at the input of the comparator in each REFYDRVS 116S. For eachreference cell and data cell in the page being programmed, the cellvoltage VCELLD(L) or VCELLR(L) is read. Depending on the output B[0:3]of the data latches (a) for each REFYDRVS 116S the appropriate programverify voltage VREFR(L) is compared to the reference cell read backvoltage VCELLR(L) and (b) for each YDRVS 110S, SYDRVS 114S, RYDRVS 112S,the appropriate program verify voltage VREFD(L) is compared with datacell read back voltage VCELLD(L) to indicate whether further programmingis required. If no further programming is required for a particularreference cell or data cell, it is put in the program inhibit mode. Ifthe Program Pulse Count=MAXPC is not true, then the cells are placed inthe program mode and another programming pulse is applied to all thecells in the page, including the reference cells. Cells which are in theprogram inhibit mode do not get any additional programming. Cells whichare not in the program inhibit mode get additional programming. Afterthe programming pulse is applied, the program pulse count is incrementedand the cells are placed in the voltage-mode read to verify if furtherprogramming is required. This iterative verify-program loop is continueduntil either all the cells in the page including the reference cells arein the program inhibit mode or when the program pulse count=MAXPC istrue. If program pulse count=MAXPC true condition is reached, before allcells in the page including the reference cells are all in programinhibit mode, then the program flag=fail condition is set, BUSY signalis reset and the programming cycle is done. Whenever the All Cells inProgram Inhibit Mode=true condition is reached, the flow moves to thenext step as shown in FIG. 22A.

As shown in FIG. 22A, next, for each level L, upper program marginverify voltage UMV(L)=VCELLR(L)−VUM(L) is generated, where VUM(L) is theupper margin voltage for level L. Depending on the data latch outputB[0:3] of the data latches in the respective YDRVS 110S, SYDRVS 114S,RYDRVS 112S the appropriate voltage UMV(L) is compared with read backcell voltage VCELLD(L) for all the data cells. If the result ofcomparison indicates that all upper cell margins are not within limitsthen a program flag=fail condition is set; BUSY signal is reset andprogramming cycle is done. If the result of comparison indicates thatall the upper cell margins are within limits then a program flag=failcondition is not set and then, for each level L, lower program marginverify voltage LMV(L)=VCELLR(L−1)+VLM(L) is generated, where VLM(L) isthe lower margin voltage for level L. Depending on the data latch outputB[0:3] of the data latches in the respective YDRVS 110S, SYDRVS 114S,RYDRVS 112S the appropriate voltage LMV(L) is compared with read backcell voltage VCELLD(L). If the result of comparison indicates that alllower cell margins are not within limits then a program flag=failcondition is set; BUSY signal is reset and programming cycle is done. Ifthe result of comparison indicates that all the lower cell margins arewithin limits then a program flag=fail condition is not set and BUSYsignal is reset and programming cycle is done. The program flag=failindicates the programming cycle has been unsuccessful to program thecurrent page. It does not indicate specifically which cell or cellscaused the unsuccessful programming.

Page Read Cycle

FIG. 23 shows the flow diagram for the page read cycle. During a pageread cycle a plurality of memory cells are read in parallel. Howeverthis algorithm is equally applicable for single cell read. After thepage read command is issued along with the address of the page to beread, the BUSY signal is set, RESTOREL and RESTOREH flags are reset, thedata latches in the YDRVS 110S, SYDRVS 114S, RYDRVS 112S are set tooutput B[0:3]=1111 and N is set to 3. N represents the number of bitsstored per memory cell. In another embodiment a configuration (fuse) bitinitialization is executed to load in data from fuse non-volatile memorycells to volatile latches located in the fuse circuit block (FUSECKT)182 at this step. All the cells in the addressed page are placed in thevoltage-mode read and the cell voltages, VCELLR(L) for reference cellsand VCELLD(L) for data cells are read. BN is forced to “0” and the readverify voltage VCELLR(L), which is one of the reference read backvoltages dependent on B3, B2, B1, B0, is compared with the cell readback voltage VCELLD(L). For each cell, if the VCELLD(L)>VCELLR(L) thenBN is latched as “1”, otherwise BN is latched as “0”. The loop continuesuntil all the bits B3, B2, BE1, B0 are latched and N=0. Next, as shownin FIG. 24, for each level L, a MARGIN RESTORE LOW VoltageVRSTRL(L)=VCELLR(L)−VRSTL(L) is generated, where VRSTL(L) is the restorelow margin voltage. Depending on the latched bits B3, B2, B1, B0 on eachof the YDRVS 110S, SYDRVS 114S, RYDRVS 112S, the voltage VRSTRL(L) iscompared with the respective data cell read back voltage VCELLD(L). IfVCELLD(L)>VRSTRL(L) for any one of the cells, then the RESTOREL flag isset. Next, for each level L a MARGIN RESTORE HIGH VoltageVRSTRH(L)=VCELLR(L−1)+VRSTH(L) is generated, where VRSTH(L) is therestore high margin voltage. Depending on the latched bits B3, B2, B1,B0 on each of the YDRVS 110S, SYDRVS 114S, RYDRVS 112S, the voltageVRSTRH(L) is compared with the respective data cell read back voltageVCELLD(L). If VCELLD(L)<VRSTRH(L) for any one of the cells, then theRESTOREH flag is set, otherwise RESTOREH flag is not set. Next, as shownin FIG. 25, BUSY signal is reset and the byte count ND is initialized toNDI. NDI is the byte count of the existing byte address location. Allbits in the respective YDRVSs, SYDRVSs, or RYDRVSs data latches areready to be sequentially read. Whenever READ CL0CK=Y, the RED_ADD_TRUEis checked for that byte address location. If RED_ADD_TRUE=Y, then datafrom RYDRVS 112S is output to the IO port IO[0:7] 1001, otherwise datafrom YDRVS 110S is output to the io port IO[0:7] 1001. If READ CL0CK=Nand ENABLE=Y then the flow loops back until READ CLOCK=Y or ENABLE=N.After all the data is output i.e. ND>MAXND=Y or if ENABLE=N, the Pageread cycle is done. If ND>MAXND is=N, then ND is incremented and theflow loops back to check the READ CL0CK.

FIG. 26 shows the details of an embodiment of YDRVS 110S. SYDRVS 114Sand RYDRVS 112S have similar details. The description given for YDRVS110S is equally applicable for SYDRVS 114S and RYDRVS 112S. In thisembodiment 4 bits are stored per memory cell, hence four data latchesare required per YDRVS 110S. A set of four data latches (DATALAT3) 10,(DATALAT2) 11, (DATALAT1) 12, (DATALAT0) 13 holds the data during theDATAIN step of a page programming cycle or holds the data during a LATCHEN=1 or=0 step during a page read cycle. Data is loaded into DATALAT310, DATALAT2 11, DATALAT1 12, DATALAT0 13 through the DIN3 14, DIN2 15,DIN1 16, DIN0 17 lines respectively and read out from the DATALAT3 10,DATALAT2 11, DATALAT1 12, DATALAT0 13 through the DOUT3 18, DOUT2 19,DOUT1 20, DOUT0 21 lines respectively. Lines (DIN3) 14, (DIN2) 15,(DIN1) 16, (DIN0) 17, (DOUT3) 18, (DOUT2) 19, (DOUT1) 20, (DOUT0) 21connect to BYTESEL 140 for YDRV 110 and connect to blocks 144, 142 forSYDRV 114, RDYRV 112 respectively. During page program cycle, lines (B3)22, (B2) 23, (B1) 24, (B0) 25 are outputs of DATALAT3 10, DATALAT2 11,DATAL1 12, DATALAT0 13, respectively, and have a latched logicalrelationship to the lines (DIN3) 14, (DIN2) 15, (DIN1) 16, (DIN0) 17,respectively. During page read cycle lines B3 22, B2 23, B1 24, B0 25are output of DATALAT3 10, DATALAT2 11, DATALAT1 12, DATALAT0 13respectively and represent the 4 bits read out of the cell. Depending onthe status of lines (B3) 22, (B2) 23, (B1) 24, and (B0) 25, theREFERENCE MULTIPLEXER 26 couples one of the lines VR0 through VR15 toone input of the VOLTAGE COMPARATOR 27. The output of the VOLTAGECOMPARATOR 27 connects to the input of the LATCH 28. Under the controlof ALGOCNTRL 164, the line ENLATCOMP 29 functions as a strobe signal toenable the LATCH 28 during a certain time to latch the output of theVOLTAGE COMPARATOR 27. Line RBYLATCOMP 30 resets the LATCH 28 atsuitable times under the control of ALGOCNTRL 164. The PROGRAM/READCONTROL 31 outputs lines COMPOR 32 and COMPORB 33. COMPOR 32 and COMPORB33 lines are connected together in a wire-OR manner for all YDRV 110,SYDRV 114, and RYDRV 112. The PROGRAM/PROGRAM INHIBIT SWITCH 34 puts thememory cell coupled to it indirectly through line BLIN 35 into a programor program inhibit mode under the control of PROGRAM/READ CONTROL 31.Line BLIN 35 goes to the PSEL 120 for YDRV 110 and to blocks 124, 122for SYDRV 114, RYDRV 112 respectively. The lines VR0 through VR15individually are coupled to the output of a VRBUFFER 750.

FIG. 27 shows the details of a LATCH 28 block, a PROGRAM/READ CONTROL 31block and a PROGRAM/PROGRAM INHIBIT 34 block. The VROUT line 55 couplesthe output of REFERENCE MULTIPLEXER 26 to the positive input of aVOLTAGE COMPARATOR 27. The line COMPOUT 58 couples the output of theVOLTAGE COMPARATOR 27 to the D input of a latch 59. ENLATCOMP 29 goes tothe EN input of the latch 59. ENLATCOMP 29 acts as a strobe signal forthe latch. When ENLATCOMP 29 is at logic high the latch 59 outputs thelogic level on D input to the Q output. QB is the inverted logic levelof Q. When ENLATCOMP 29 goes to logic low, the latch 59 latches thelogic level on D input. RBYLATCOMP 30 goes to the reset R input of thelatch 59. When RBYLATCOMP 30 is logic low latch 59 is reset, whereby Qis at logic low and QB is at logic high. Line COMLATQ 40 couples the Qoutput of the latch 59 to the gate of a NMOS transistor N1 43. LineCOMLATQB 41 couples the QB output of the latch 59 to the gate of a NMOStransistor N2 44. Line COMLATQ 40 also couples to the data latchesDATALAT3 10, DATALAT2 11, DATAL1 12, DATALAT0 13. COMLATQ 40 alsocouples to one input of a 2 input NAND gate NAND 49. The other input ofthe NAND 49 is coupled to the signal READ2B. READ2B is at logic highduring page programming cycle and at logic low during page read cycle.The line NDO 52 couples the output of NAND 49 to the input of aninverter INV 48 and also to the gate inputs of PMOS transistor P1 45 andNMOS transistor 1N3 47. The line INVO 53 couples the output of INV 48 tothe gate of a PMOS transistor P2 46. Line BLIN 35 connects to oneterminal of each of P1 45, N3 47 and P2 46. BLIN 35 also connects to thenegative input of VOLTAGE COMPARATOR 27. The other terminal of P1 45 isconnected to inhibit voltage input VIH 57. Line N4D 54 connects theother terminals of N3 47 and P2 46 to one terminal of NMOS transistor N450. Line N5D 60 connects the other terminal of N4 50 to one terminal ofNMOS transistor N5 51. The other terminal of N5 51 is connected toground. The gates of N4 50 and N5 51 are connected to inputs VBIYDRVCAS56 and VBIYDRV 57 respectively. N4 50 and N5 51 form a current biascircuit whereby a constant current load is placed on the BLIN 35 whenINVO 53 is at logic low and NDO 52 is at logic high. N4 50 and N5 51together represent the predetermined bias current for the voltage modesensing as shown in FIG. 2C.

After the page program command and the address of the page to be programis issued, the data to be programmed is loaded in the data latchesDATALAT3 10, DATALAT2 11, DATAL1 12, DATALAT0 13 of each of the YDRVS110S, SYDRVS 114S or RYDRVS 112S. The REFERENCE MULTIPLEXER 26 thencouples one of the inputs VR0 through VR15 to its output VROUT 55.During a program verify cycle VREFD(0) through VREFD(15) are availableon the VR0 through VR15 lines respectively. VR0 through VR15 arecommonly coupled to REFERENCE MULTIPLEXER 26 of all the YDRV 110, SYDRV112, RYDRV 14. The REFYDRVS 116S have the data latches internally set.In this embodiment there are 16 REFYDRVS 116S. Each REFYDRVS 116S isused for a specific level. For example, the data latches of a REFYDRVS116S used for level 5 will be internally set to program level 5 intoreference cells coupled to it. VR0 through VR15 are commonly coupled toREFERENCE MULTIPLEXER 26 of all the REFYDRVS 116S. During a programverify cycle, VREFR(0) through VREFR(15) are respectively available atthe VR0 through VR15 lines of a REFYDRVS 116S. Depending on the outputB3, B2, B1, B0 of the data latches DATALAT3 10, DATALAT2 11, DATALAT112, DATALAT0 13 within each YDRVS 110S, SYDRVS 114S, SYDRVS 112S onespecific voltage VREFD(0) through VREFD(15) is output to the input ofthe VOLTAGE COMPARATOR 27. Depending on the output B3, B2, B1, B0 of thedata latches DATALAT3 10, DATALAT2 11, DATALAT1 12, DATALAT0 13 withineach REFYDRV 116 one specific voltage VREFR(0) through VREFR(15) isoutput to the input of the VOLTAGE COMPARATOR 27.

The latch 59 within each REFYDRVS 116S, YDRVS 110S, SYDRVS 114S andRYDRVS 112S are all reset by pulsing line RBYLATCOMP 30. RBYLATCOMP 30is commonly connected to the reset input of the latch 59 within eachREFYDRVS 116S, YDRVS 110S, SYDRVS 114S, and RYDRVS 112S. After latch 59is reset, COMLATQ 40 is at logic low. The NAND 49 then outputs logichigh to line NDO 52. Output of INV 48 then is at logic low on line INVO53. With NDO 52 at logic high and INVO 53 at logic low transistors N3 47and P 246 couple BLIN 35 to N4 50. P1 45 de-couples the inhibit voltageVIH 57 from BLIN 35. The memory cell is placed in the voltage read modeand the cell read back voltage VCELLR(L) or VCELLD(L) is available onBLIN 35. At this point, the VOLTAGE COMPARATOR 27 compares the voltagesat its inputs. If voltage on BLIN 35 is higher then voltage on VROUT 55the output COMPOUT 58 is low, otherwise it is high. At this time apositive going strobe pulse is applied to the ENLATCOMP 29 common to allthe latches 59 in REFYDRVS 116S, YDRVS 110S, SYDRVS 114S and RYDRVS112S, to latch the status of line COMPOUT 58. If COMPOUT 58 is low, thenthe COMLATQ 40 remains at logic low.

If COMPOUT 58 is high, then the COMLATQ 40 switches to logic high. Ifduring an iteration of verify-program cycles any one of the latches 59latches a logic high on COMLATQ 40, called a program inhibit state, thenfor that specific REFYDRVS 116S, YDRVS 110S, SYDRVS 114S or RYDRVS 112S,the line NDO 52 is at low and the line INVO 53 is at logic high. Withlatch 59 in a program inhibit state, BLIN 35 is de-coupled from N4D 54and there is no current load, whereas, BLIN 35 is coupled to the inhibitvoltage VIH 57 through P1 45. With latch 59 in the program inhibitstate, further programming pulses do not cause programming.

The line COMPOR 32 is connected in a wire-OR fashion to all the COMPOR32 lines of each REFYDRVS 116S, YDRVS 110S, SYDRVS 114S or RYDRVS 112S.There is a pull up load coupling the COMPOR 32 line to the power supply.Similarly, the line COMPORB 33 is connected in a wire-OR fashion to allthe COMPORB 33 lines of each REFYDRVS 116S, DRVS 110S, SYDRVS 114S orRYDRVS 112S. There is a pull up load coupling the COMPORB 33 line to thepower supply. The COMPORB line 33 goes high whenever all the latches 59have reached the program inhibit mode. When the Program PulseCount=MAXPC is reached, the ALGOCNTRL 164 latches the status of COMPORBline 33 in a status latch in block INPUT LOGIC 160. The status latch canbe read at one of the IO[0:7] 1001 lines by the external host. IfALGOCNTRL 164 latches a logic low in the status latch in block INPUTLOGIC 160 then a program fail condition is reached and the ALGOCNTRL 164goes out of the page programming cycle.

If at the end of any verify-program iteration, the COMPOR 32 line goeshigh, the ALGOCNTRL 164 sequences to the margin verify mode. All latches59 are reset. All cells are placed in the voltage read mode by READB 52at logic low. At this time inhibit voltage is de-coupled from BLIN 35and current bias transistor N4 50 is coupled to BLIN 35. Cell voltagesVCELLR(L) or VCELLD(L) are respectively available on BLIN 35 of aREFYDRVS 116S or BLIN 35 of YDRVS 110S, SYDRVS 114S, or RYDRVS 112S.During program margin verify the voltages read back from the data cellsare checked for adequate margin from voltages read back from referencecells for each programmed level L. In the Upper Program Margin Verifymode, voltages UMV(0) through UMV(15) are placed on the VR0 throughVR(15). Depending on the output B3, B2, B1, B0 of the data latchesDATALAT3 10, DATALAT2 11, DATALAT1 12, DATALAT0 13 within each YDRVS110S, SYDRVS 114S, RYDRVS 112S one specific voltage UMV(O) throughUMV(15) is output to the input VROUT 55 of the VOLTAGE COMPARATOR 27. Atthis time the VOLTAGE COMPARATOR 27 compares the voltages at its inputs.If voltage on BLIN 35 is higher then voltage on VROUT 55 the outputCOMPOUT 58 is low, otherwise it is high. At this time a positive goingstrobe pulse is applied to the ENLATCOMP 29 common to all the latches 59in YDRVS 110S, SYDRVS 114S and RYDRVS 112S, to latch the status of lineCOMPOUT 58. If COMPOUT 58 is low, then the COMLATQ 40 remains at logiclow. If COMPOUT 58 is high, then the COMLATQ 40 switches to logic high.At this time, if LGOCNTRL 164 latches a logic low in the status latch inINPUT LOGIC 160 block by looking at the status of the COMPORB 33 line,then a program fail condition is reached and the ALGOCNTRL 164 goes outof the page programming cycle. Otherwise, ALGOCNTRL 164 sequences to theLower Program Margin Verify mode.

In the Lower Program Margin Verify mode, all latches 59 are reset.Voltages LMV(0) through LMV(15) are placed on the VR0 through VR(15).Depending on the output B3, B2, B1, B0 of the data latches (DATALAT3)10, (DATALAT2) 11, (DATALAT1) 12, (DATALAT0) 13 within each YDRVS 110S,SYDRVS 114S, RYDRVS 112S one specific voltage LMV(0) through LMV(15) isoutput to the input VROUT 55 of the VOLTAGE COMPARATOR 27. At this timethe VOLTAGE COMPARATOR 27 compares the voltages at its inputs. Ifvoltage on BLIN 55 is higher then voltage on VROUT 55 the output COMPOUT58 is low, otherwise is high. At this time a positive going strobe pulseis applied to the ENLATCOMP 29 common to all the latches 59 in YDRVS110S, SYDRVS 114S and RYDRVS 112S, to latch the status on line COMPOUT58. If COMPOUT 58 is low, then the COMLATQ 40 remains at logic low. IfCOMPOUT 58 is high, then the COMLATQ 40 switches to logic high. At thistime, if ALGOCNTRL 164 latches a logic low in the status latch in INPUTLOGIC 160 block by looking at the status of the COMPOR line 32, then aprogram fail condition is reached and the ALGOCNTRL 164 goes out of thepage programming cycle.

During page read cycle, after page read command and the page address isissued, the reference and the data cells are placed in the voltage readmode. At this time all the B3[0:3] lines output 1111. VR0 through VR15have VCELLR(0) through VCELLR(15). VCELLR(0) through VCELLR(15) are thevoltages read out of the reference cells of the page being read. Underthe control of the ALGOCNTRL 164 block 4 bits are sequentially read intothe data latches (DATALAT3) 10, (DATALAT2) 11, (DATALAT1) 12, (DATALAT0)13. For example, B3 is read by forcing the output of DATALAT3 to outputB3=0. At this time B[0:3]=1110. The REFERENCE MULTIPLEXER 26 thenoutputs VCELLR(7) on the VROUT 55 in each of the YDRVS 110S, SYDRVS 114Sand RYDRVS 112S. The output COMPOUT 58 of the VOLTAGE COMPARATOR 27 ishigh or low depending on whether voltage VCELLD(L) on the BLIN 35 islower or higher relative to voltage VCELLR(7) on line VROUT 55. IfCOMPOUT 58 is high then a logic high is latched into DATALAT3 10 andB3=0, otherwise logic low is latched and B3=1. Next, B2 is read byforcing the output of DATALAT2 11 to output B2=0. At this timeB[0:3]=101B3. B3 is the output of DATALAT3 10 from previous sequence.The REFERENCE MULTIPLEXER 26 then outputs VCELLR(L), depending on 110B3on the VROUT 55 line in each of the YDRVS 110S, SYDRVS 114S and RYDRVS112S. The output COMPOUT 58 of the VOLTAGE COMPARATOR 27 is high or lowdepending on whether voltage VCELLD(L) on the BLIN 35 is lower or higherrelative to voltage VRCELL(L) on line VROUT 55. If COMPOUT 58 is highthen a logic high is latched into DATALAT2 11 and B2=0, otherwise logiclow is latched and B2=1. In this manner, the next two sequences latchtwo bits into the DATALAT1 12 and DATALAT0 13.

After all 4 bit from the cell are latched into the DATALAT3 10, DATALAT211, DATAL1 12, DATALAT0 13 for all the YDRVS 110S, SYDRVS 114S andRYDRVS 112S, the restore margins are checked. All latches 59 are reset.First the RESTORE LOW margin is checked. At this time, for each level 0through 15, MARGIN RESTORE LOW Voltage VRSTRL(0) through VRSTRL(15) isplaced at the VR0 through VR15 lines respectively. Depending on eachoutputs B3, B2, B1, B0 of the data latches DATALAT3 10, DATALAT2 11,DATALAT1 12, DATALAT0 13 within each YDRVS 110S, SYDRVS 114S and RYDRVS112S, the REFERENCE MULTIPLEXER 26 outputs one of VRSTRL(0) throughVRSTRL(15) on line VROUT 55 going into the positive input of the VOLTAGECOMPARATOR 27. ENLATCOMP 29 is strobed with the positive pulse to latchthe status of the COMPOUT 58 line. If data cell read out voltageVCELLD(L) on BLIN 35 line is higher than voltage VRSTRL(L) on VROUT 55line then COMLATQ 40 remains at logic low and COMLATQB 41 at logic high.Otherwise, COMLAT 40 is at logic high and COMLATQB 41 at logic low. Atthis time, if ALGOCNTRL 164 latches a logic low in the RESTORE LOW latchin INPUT LOGIC 160 block by looking at the status of the COMPORB line33, then a restore low flag condition is reached. Next, all latches 59are reset.

Next the RESTORE HIGH margin is checked. At this time, for each level 0through 15, MARGIN RESTORE HIGH Voltage VRSTRH(0) through VRSTRH(15) isplaced at the VR0 through VR15 lines respectively. Depending on eachoutputs B3, B2, B1, B0 of the data latches DATALAT3 10, DATALAT2 11,DATALAT1 12, DATALAT0 13 within each YDRVS 110S, SYDRVS 114S and RYDRVS112S, the REFERENCE MULTIPLEXER 26 outputs one of VRSTRH(0) throughVRSTRH(15) on line VROUT 55 going into the positive input of the VOLTAGECOMPARATOR 27. ENLATCOMP 29 is strobed with the positive pulse to latchthe status of the COMPOUT 58 line. If data cell read out voltageVCELLD(L) on BLIN 35 line is higher than voltage VRSTRH(L) on VROUT 55line then COMLATQ 40 remains at logic low and COMLATQB 41 at logic high.Otherwise, COMLAT 40 is at logic high and COMLATQB 41 at logic low. Atthis time, if ALGOCNTRL 164 latches a logic low in the RESTORE HIGHlatch in INPUT LOGIC 160 block by looking at the status of the COMPORline 32, then a restore high flag condition is reached.

At this time, 4 bits from every cell with the page being read arelatched into the respective data latches within each YDRVS 110S, SYDRVS114S and RYDRVS 112S. Next under the control of the READ CLOCK data issequentially read on IO[0:7]. If after READ CLOCK the RED_ADD_TRUE=Ycondition is true then the data is read from the addressed RYDRVS 112Sotherwise data is read from the addressed YDRVS 110S or SYDRVS 114S.

FIG. 19B shows various voltages generated and used in another embodimentof the current invention for program verifying, program marginverifying, read sensing and restore high or low margin verifying. Inthis embodiment the program margin verify voltage VREFR(L)-VRM(L) andVREFD(L)-DM(L) for a level L of a reference cell and a data cellrespectively, are generated by the block V&IREF 172 independent of thevoltages VCELLR(L) and VCELLD(L) programmed into the reference cell anddata cell respectively. The voltage VRM(L) for a level L of thereference cells can be unique for each level or the same for all levels.The voltage VDM(L) for a level L of the data cells can be unique foreach level or the same for all levels.

FIG. 22B shows the portion of the flow for the page programming cyclethat uses the voltages as shown in FIG. 19B. In the flow shown in FIG.22B, only one program margin verify comparison is made instead of two asshown in FIG. 22A. This has the advantage of reducing the total time forcompletion of a page programming cycle.

FIG. 22C shows an alternative embodiment of the flow shown in FIG. 22B.At the end of the programming, a BSERV operation is done to verify thatthe read operation is operational versus the data in. The BSERVoperation is a binary search read verification operation that issubstantially the same as described in FIGS. 23 and 24 with theadditional step of comparing resulting digital bits BR<3:0> from thebinary search with a stored digital bits B<3:0> from loading data in. Ifthe comparison is not true, the program flag is set to indicate programfailure. The operation further ensures that all cells are within anoperational range, for example not out of range due to programmingovershoot to the next levels.

The embodiment shown in FIGS. 19B and 22B can be used in combinationwith the embodiment shown in FIGS. 19A and 22A. As discussed in themultilevel reference system section above, the embodiment shown in FIGS.19B and 22B can be used when a selected page programs for the first timeafter block erase. For subsequent page programming cycles on the samepage, the embodiment shown in FIGS. 19A and 22A is advantageous sincethe VCELLR(L) values may shift between initial page programming andsubsequent page programming.

FIG. 28 is a block diagram illustrating a memory system 2800 for amultilevel memory.

The memory system 2800 comprises a plurality of memory arrays 2801arranged in rows and columns of memory arrays 2801. Each memory array2801 comprises a plurality of memory subarrays 2802, a plurality oflocal sense amplifiers 2804, and a plurality of global sense amplifiers2806. In one embodiment, a local sense amplifier 2804 is disposedadjacent to a memory subarray 2802. In another embodiment, the localsense amplifier 2804 is shared between a plurality of memory subarrays2802. The local sense amplifier 2804 reads the contents of the memorycells within the corresponding memory subarray 2802. The memorysubarrays 2802 are arranged in rows and columns. The local senseamplifiers 2804 coupled to a column of memory subarrays 2802 are coupledto a global sense amplifier 2806. The memory cells may include redundantcells, reference cells or spare cells.

FIG. 29A is a block diagram illustrating an inverter mode sensingcircuit 2900.

The inverter mode sensing circuit 2900 comprises a PMOS transistor 2902,a plurality of NMOS transistors 2904 and 2906, a feedback circuit 2908,a plurality of memory cells 2910, and a comparator 2912. For clarity,only one memory cell 2910 and one NMOS transistor 2906 are shown for asubarray, but the subarray comprises a plurality of memory cells 2910arranged in columns. Each column has a corresponding NMOS transistor2906 or a plurality of NMOS transistors 2906 arranged in series. Onlyone column with one memory cell 2910 is shown.

The comparator 2912 determines the voltage of the memory cell bycomparing the cell voltage (VCELL) 2914 to a reference voltage (VREF)2916 in a manner described above. The PMOS transistor 2902, the NMOStransistors 2904 and 2906 and the memory cells 2910 are coupled inseries between the supply voltage and ground. The selected memory cell2910 is read by applying a control gate reference voltage (VCGRD) 2917on the control gate of the memory cell 2910. The column of memory cells2910 and an associated bit line has a capacitance 2918 that slows thesensing of the memory cells 2910. The NMOS transistor 2906 functions asa switch to couple the column of memory cells 2910 to the sensingportion of the circuit. The feedback circuit 2908 controls biasing ofthe NMOS transistor 2904 to stabilize the cell voltage 2914. The drainof the diode connected PMOS transistor 2902 is coupled to the cellvoltage 2914. Inverter mode sensing may also be referred to as currentmode sensing or common source sensing. In another embodiment of currentmode sensing, the source line (CL) (as shown coupled to ground) and thebitline are interchanged, and thus the voltage on the source line iscoupled to the readout voltage 2914. In this case, the arrayarchitecture uses only one source line in read at a given time, forexample by multiplexing through decoder circuitry or over time. Thismode is to be known as Inverse Current Mode sensing.

FIG. 29B is a block diagram illustrating a voltage mode sensing circuit2950.

The voltage sensing circuit 2950 is similar to the inverter mode sensingcircuit 2900 except that a current source 2952 replaces the PMOStransistor 2902 and is coupled to ground, the memory cell 2910 iscoupled to a reference bias, and the NMOS transistor 2904 and thefeedback circuit 2908 are omitted. The voltage mode sensing may also bereferred to as source follower sensing.

FIG. 30 is a block diagram illustrating a wide range, high speed voltagemode sensing circuit 3000.

The memory array 2800 includes a plurality of voltage mode sensingcircuits 3000. The voltage mode sensing circuit 3000 comprises a PMOStransistor 3002, a plurality of NMOS transistors 3004, 3006, 3007, afeedback circuit 3008, a plurality of memory cells 3010, a currentsource (IRCELL) 3011, and a comparator 3012. For clarity, only onememory cell 3010, one NMOS transistor 3006, and one NMOS transistor 3007are shown for a subarray, but the subarray comprises a plurality ofmemory cells 3010 arranged in columns. Each column has a correspondingNMOS transistor 3006. Only one column with one memory cell 3010 isshown. Possible decoding circuitry between the current source 3011 andthe memory cell 3010 and between the current source 3011 and the NMOStransistor 3007 is not shown.

The comparator 3012 determines the voltage of the memory cell bycomparing a cell voltage (VCELL) 3014 to a reference voltage (VREF) 3016in a manner described above. The PMOS transistor 3002, the NMOStransistors 3004, 3006 and 3007 are coupled in series between the supplyvoltage and ground. The current source 3011 is coupled between the gateof the NMOS transistor 3002 and ground. The memory cell 3010 is coupledbetween a reference voltage (VCLRD) and the common node formed of thecurrent source 3011 and the gate of the NMOS transistor 3007.

The selected memory cell 3010 is read by applying a control gatereference voltage (VCGRD) 3017 on the control gate of the memory cell3010. The biasing of the gate of the NMOS transistor 3007 by the currentsource 3011 and the memory cell 3010 controls the voltage on the bitline.

The NMOS transistor 3006 functions as a switch to couple the column ofNMOS transistors 3007 and the associated memory cells 3010 to thesensing portion of the circuit. The feedback circuit 3008 controlsbiasing of the NMOS transistor 3004 to stabilize the cell voltage 3014.The drain of the diode connected PMOS transistor 3002 is coupled to thecell voltage 3014.

FIG. 31 is a block diagram illustrating a voltage mode sensing circuit3100.

The voltage mode sensing circuit 3100 comprises a plurality of memorysubarrays 3150, a plurality of local sense amplifiers 3152, and aplurality of global sense amplifiers 3154. The local sense amplifier3152 includes a local source follower stage. The global sense amplifier3154 includes a common source stage.

The memory array 3150 includes columns of memory cells 3110 coupled tofirst bitlines 3151.

Each local sense amplifier 3152 is coupled to a memory subarray 3150. Inone embodiment, the local sense amplifier 3152 is disposed adjacent thememory subarray 3150. The local sense amplifier 3152 includes aselection circuit 3153 that couples a selected bitline 3151 to a bitline3155. In one embodiment, the selection circuit 3153 comprisestransistors. The local sense amplifier 3152 senses the selected memorycell on the bitline 3151 and provides a voltage to a global senseamplifier 3154.

The local sense amplifier 3152 comprises an NMOS transistor 3107 coupledbetween the bitline 3155 and ground, and includes a gate coupled to thebitline 3151. A current source 3111 is coupled between the gate of theNMOS transistor 3107 and ground.

The global sense amplifier 3154 comprises a comparator 3112, a PMOStransistor 3102 and a selection circuit 3158. The selection circuit 3158couples the selected one of the bitlines 3155 to a common node formed ofa voltage cell input 3114 of the comparator 3112 and the drain of thediode connected PMOS transistor 3102. A reference voltage 3116 isapplied to the second input of the comparator 3112.

The local sense amplifier 3152 provides a larger voltage range by usingoptimally low current bias. The global sense amplifier 3154 includes acommon source stage with a PMOS transistor 3114 as a load, and buffersthe column capacitance.

The voltage mode sensing circuit 3100 further comprises a plurality ofreference subarrays 3170, a plurality of local sense amplifiers 3172,and a plurality of global sense amplifiers 3174. The reference subarrays3170 comprise a plurality of reference cells for storing referencesignals. In one embodiment, the reference subarrays 3170 are similar tothe memory subarrays 3150. The local sense amplifiers 3172 read thereference subarrays 3170. In one embodiment, the local sense amplifiers3172 are similar to the local sense amplifiers 3152. The global senseamplifiers 3174 detect and amplify the voltage from the local senseamplifiers 3172.

The global sense amplifier 3174 comprises a comparator 3173, a PMOStransistor 3176 and a selection circuit 3178, which are arranged insimilar manner as the comparator 3112, the PMOS transistor 3102 and theselection circuit 3158 of the global sense amplifier 3154, except thecomparator 3173 is configured as a buffer. The comparator 3173 serves asa comparator in sensing the reference cells and serves as a buffer fordriving the reference level.

FIG. 32 is a block diagram illustrating a voltage mode sensing circuit3200.

The voltage mode sensing circuit 3200 includes like elements as thevoltage mode sensing circuit 3100 (FIG. 31) and are given like referencenumbers. The voltage mode sensing circuit 3200 comprises a memory array3150, a plurality of local sense amplifiers 3252 and a plurality ofglobal sense amplifiers 3254. The local sense amplifier 3252 includes alocal source follower stage and includes a PMOS source follower as partof the global sense amplifier. The global sense amplifier 3254 includesa source follower stage.

Each local sense amplifier 3252 is coupled to a memory subarray 3150. Inone embodiment, the local sense amplifier 3252 is disposed adjacent thememory subarray 3150. The local sense amplifier 3252 includes aselection circuit 3253 that couples a selected bitline 3151 to a bitline3255. In one embodiment, the selection circuit 3253 comprisestransistors. The local sense amplifier 3252 senses the selected memorycell on the bitline 3151 and provides a voltage to a global senseamplifier 3254.

The local sense amplifier 3252 comprises a PMOS transistor 3207 coupledbetween the bitline 3255 and ground, and includes a gate coupled to thebitline 3151. A current source 3211 is coupled between the gate of thePMOS transistor 3207 and ground. The local sense amplifier 3252 providesa maximum voltage range by using low current bias.

The global sense amplifier 3254 comprises a comparator 3212, a currentsource 3202 and a selection circuit 3258. The current source 3202couples the supply voltage to the cell voltage terminal 3214 of thecomparator 3212 to ground. The selection circuit 3258 couples theselected one of the bitlines 3255 to a common node formed of a voltagecell input 3214 of the comparator 3212 and the current source 3202. Areference voltage 3216 is applied to the second input of the comparator3212.

The global sense amplifier 3254 buffers the column capacitance.

The voltage mode sensing circuit 3200 further comprises a plurality ofreference subarrays 3170, a plurality of local sense amplifiers 3282,and a plurality of global sense amplifiers 3274. The reference subarrays3170 comprise a plurality of reference cells for storing referencesignals. In one embodiment, the reference subarrays 3170 are similar tothe memory subarrays 3150. The local sense amplifiers 3282 read thereference subarrays 3170. In one embodiment, the local sense amplifiers3282 are similar to the local sense amplifiers 3252. The global senseamplifiers 3274 detect and amplify the voltage from the local senseamplifiers 3282.

The global sense amplifier 3274 comprises a comparator 3292, a currentsource 3272 and a selection circuit 3278, which are arranged in similarmanner as the comparator 3212, the current source 3202 and the selectioncircuit 3258 of the global sense amplifier 3254, except the comparator3292 is configured as a buffer. The comparator 3292 serves as acomparator in sensing the reference cells and serves as a buffer fordriving the reference level.

FIG. 33 is a block diagram illustrating voltage mode sensing circuit3300.

The voltage mode sensing circuit 3300 includes like elements as thevoltage mode sensing circuit 3200 (FIG. 32) and are given like referencenumbers. The voltage mode sensing circuit 3300 comprises a memory array3150, a plurality of local sense amplifiers 3352 and a plurality ofglobal sense amplifiers 3354. The local sense amplifier 3352 includes alocal source follower stage and includes an NMOS source follower as partof the global sense amplifier. The global sense amplifier 3354 includesa source follower stage.

Each local sense amplifier 3352 is coupled to a memory subarray 3150. Inone embodiment, the local sense amplifier 3352 is disposed adjacent thememory subarray 3150. The local sense amplifier 3352 includes aselection circuit 3253 that couples a selected bitline 3151 to a bitline3355. In one embodiment, the selection circuit 3253 comprisestransistors. The local sense amplifier 3252 senses the selected memorycell on the bitline 3151 and provides a voltage to a global senseamplifier 3254.

The local sense amplifier 3352 comprises an NMOS transistor 3307 coupledbetween the bitline 3355 and a supply voltage terminal, and includes agate coupled to the bitline 3151. A current source 3311 is coupledbetween the gate of the NMOS transistor 3307 and ground. The local senseamplifier 3252 provides a maximum voltage range by using low currentbias.

The global sense amplifier 3354 comprises a comparator 3312, a currentsource 3302 and a selection circuit 3358. The current source 3302couples the voltage terminal 3314 of the comparator 3312 to a groundterminal. The selection circuit 3358 couples the selected one of thebitlines 3355 to a common node formed of a voltage cell input 3314 ofthe comparator 3312 and the current source 3302. A reference voltage3316 is applied to the second input of the comparator 3312. The globalsense amplifier 3354 is selectively coupled to the bitline to comparethe cell voltage to a reference voltage 3316. The global sense amplifier3354 buffers the column capacitance.

The voltage mode sensing circuit 3300 further comprises a plurality ofreference subarrays 3170, a plurality of local sense amplifiers 3382,and a plurality of global sense amplifiers 3374. The reference subarrays3170 comprise a plurality of reference cells for storing referencesignals. In one embodiment, the reference subarrays 3170 are similar tothe memory subarrays 3150. The local sense amplifiers 3382 read thereference subarrays 3170. In one embodiment, the local sense amplifiers3382 are similar to the local sense amplifiers 3352. The global senseamplifiers 3374 detect and amplify the voltage from the local senseamplifiers 3382.

The global sense amplifier 3374 comprises a comparator 3392, a currentsource 3372 and a selection circuit 3378, which are arranged in similarmanner as the comparator 3312, the current source 3302 and the selectioncircuit 3358 of the global sense amplifier 3354, except the comparator3392 is configured as a buffer. The comparator 3392 serves as acomparator in sensing the reference cells and serves as a buffer fordriving the reference level.

In another embodiment, the local sense amplifier is a common sourceamplifier, and the global sense amplifiers are NMOS source followerstages or PMOS source follower stages.

In another embodiment, the local sense amplifier is a common sourceamplifier, and the global sense amplifiers are common source amplifiers.

FIG. 34 is a block diagram illustrating a global sense amplifier 3400having an auto zeroing function.

The comparators 3012, 3112, 3212, and 3312 of FIGS. 30-33 may be theglobal sense amplifier 3400.

The sense amplifier 3400 comprises an operational amplifier 3402, a pairof capacitors 3404 and 3405, and a plurality of switches 3406 and 3407.

The capacitors 3404 and 3405 couples respective inputs 3408 and 3410 ofthe operational amplifier 3402 to the switch 3406.

In response to an auto zero (AZ) command 3416, the switches 3407selectively couples an output 3412 of the operational amplifier 3402 tothe input 3408 to equalize the voltages on the output 3412 and input3408, and selectively couples an output 3414 of the operationalamplifier 3402 to the input 3410 to equalize the output 3414 and theinput 3410. In the auto zero mode, the voltage on A terminals of thecapacitors 3404 and 3405 are set equal to the reference voltage (VREF)3418, and the B terminals of the capacitors 3404 and 3405 are equalizedto the complementary outputs of the operational amplifier 3402. Theswitch 3406 is switched by an evaluation (EVA) command 3422 to connectthe cell voltage (VCELL) 3420 to the other end of the capacitor 3405 forcomparison from the operational amplifier 3402.

The switch 3406 selectively applies the reference voltage (VREF) 3418 tothe capacitor 3404 in response to the evaluation (EVA) command 3422. Theswitch 3406 also selectively applies either the reference voltage (VREF)3418 or a cell voltage (VCELL) 3420 to the capacitor 3405 in response tothe evaluation (EVA) command 3422. The evaluation command 3422 equalizesthe signals on terminals 3404A and 3505A of the capacitors 3404 and3405.

In an alternate embodiment, the nodes 3404B and 3405B of the capacitors3404 and 3405 are reset to a fixed bias voltage. In another embodiment,the nodes 3404B and 3405B of the capacitors 3404 and 3405 are shortedtogether.

By using a capacitor for sensing, the input common load range to theoperational amplifier (or comparator) is substantially constant andindependent of the memory cell voltage or current.

FIG. 35 is a block diagram illustrating an auto zero sense amplifier3500.

The autozero sense amplifier 3500 comprises a plurality of PMOStransistors 3502 and 3504, a plurality of NZ NMOS transistors 3506 and3507, a plurality of NMOS transistors 3508 through 3516, a plurality ofcapacitors 3518 and 3519 and a plurality of transfer gates 3522 through3528.

The PMOS transistors 3502 and 3504 and the NMOS transistors 3508, 3509and 3513 and the NZ NMOS transistor 3507 are arranged as a differentialpair. The NMOS transistors 3508 and 3509 provide the differential inputpair. The NZ NMOS transistor 3507 and the NMOS transistor 3513 providebias for the NMOS transistor 3508 and 3509. The PMOS transistors 3502and 3504 are coupled for cross-coupled loading. The PMOS transistor 3502is coupled between the supply voltage and an output terminal 3530. Abias voltage 3529 is applied to the gates of the NZ NMOS transistors3506 and 3507 and the NMOS transistors 3513 and 3514.

The NMOS transistors 3510 and 3511 provide an NMOS coupled internallatch, which is active while the differential input pair is on. Thedrain of the NMOS transistor 3510 is coupled to the drain of the NMOStransistor 3509 and the gate of the NMOS transistor 3511. The drain ofthe NMOS transistor 3511 is coupled to a common node formed of the drainof the NMOS transistor 3508 and gate of the NMOS transistor 3510. The NZNMOS transistor 3506 and the NMOS transistor 3514 provide bias for theNMOS transistors 3510 and 3511 and are coupled between the common nodeformed of the sources of the NMOS transistors 3510 and 3511, and ground.

The transfer gate 3522 couples the drains of the PMOS transistors 3502and 3504 and the output 3530 to each other for equalization and quickrecovery for the next comparison in response to a release signal 3531and an inverted release signal 3532.

The capacitor 3519 couples the gate of the NMOS transistor 3509 to firstterminals of the transfer gates 3525 and 3526 which include a secondterminal coupled to a reference voltage 3534. The capacitor 3518 couplesthe gate of the NMOS transistor 3508 into first terminals of thetransfer gates 3527 and 3528, which have second terminals coupled to thereference voltage 3534 and a cell voltage 3535, respectively. Thetransfer gates 3525 and 3527 are controlled by a auto zero signal 3537and an inverted auto zero signal 3538. The transfer gates 3526 and 3528are controlled by evaluation signals 3539 and 3540.

The transfer gates 3523 and 3524 couple the drains of the PMOStransistors 3504 and 3502, respectively, to the gates of the NMOStransistors 3509 and 3508, respectively, in response to the auto zerosignal 3537 and inverted auto zero signal 3538. The NMOS transistors3512 and 3516 couple the gates of the NMOS transistors 3509 and 3508,respectively, to ground in response to a strobe signal 3542 to pull downthe transistors 3509 and 3508 to turn off the differential pair. TheNMOS transistor 3515 couples the sources of the NMOS transistors 3510and 3511 to the ground in response to the strobe signal 3542 for fulllevel latching.

The array architectures described herein may enable multilevel paralleloperation.

A pipelined read operation may be as follows. A first row is selected ina selected subarray, such as subarray 2802 or subarray 3150/3170, andthe content of selected memory cells are coupled to the local bitlineand to the global bitlines while a second row in another subarray 2802or 3150/3170 is selected and the content of the selected memory cellsare coupled to the local bitlines but not yet coupled to the globalbitlines. After the read operation completes processing the data of thefirst row, the data of the second row is enabled to couple to the globalbitlines to continue the read operation, and a third row in a differentsubarray 2802 or 3150/3170 is selected to enable the content of theselected memory cells to couple to the local bitlines but not yet to theglobal bitlines. This cycle continues until all desired data are readout. This, for example, enables continuous read of multilevel memorycells.

In another embodiment, pipelined read operation is performed byoperating on memory cells in a row in an array, such as memory array2801, while another row in another memory array 2801 is selected toenable the contents of the memory cells to be ready.

A read-while-read operation may be as follows. A read operation operateson both arrays, such as memory array 2801 (or memory subarrays 2802 or3150), simultaneously and the data are available from both arrayspossibly at the same time. In this case, for example, data latches areused to latch the data from both arrays. In another embodiment, two setsof data lines may be used to transfer the data from both arrays to anon-chip controller.

A read/write-while-write/read operation may be as follows. Similarlywhile one operation, e.g., read, is executed on an array, such assubarray 2802 or array 2801 or subarrays 3150/3170, another operation isexecuted, e.g., write, on another array such as subarray 2802 or 2801 orsubarray 3150/3170. This is possible because control circuits associatedwith decoding and sensing and/or writing may be embedded for each array.

A read/erase-while-erase/read may be as follows. Similarly while oneoperation, e.g., read, is executed on an array, such as subarray 2802 or2801 or subarray 3150/3170, another operation is executed, e.g., erase,on another array such as subarray 2802 or 2801 or subarray 3150/3170.This is possible because each array may have its own decoders andembedded control circuits associated with sensing.

An erase-while-erase operation may be as follows. Similarly while oneerase operation is executed on an array, such as subarray 2802 or 2801or subarray 3150/3170, another erase operation is executed on anotherarray, such as subarray 2802 or 2801 or subarray 3150/3170. This ispossible because each array may have its own decoders.

A write/erase-while-erase/write operation may be as follows. Similarlywhile one operation, e.g., write, is executed on an array, such assubarray 2802 or array 2801 or subarrays 3150/3170, another operation isexecuted, e.g., erase, on another array such as subarray 2802 or 2801 orsubarray 3150/3170. This is possible because each array may have its owndecoders and embedded control circuits associated with sensing and/orwriting.

A write-while-write operation may be as follows. Similarly while onewrite operation is executed on an array, such as subarray 2802 or 2801or subarray 3150/3170, another write operation is executed on anotherarray, such as subarray 2802 or 2801 or subarray 3150/3170. This ispossible because each array may have its own decoders and embeddedcontrol circuits associated with sensing and/or writing.

FIG. 36 is a block diagram illustrating a memory system 3600 for amultilevel memory including local autozero sense amplifiers and globalautozero sense amplifiers.

The memory system 3600 comprises a plurality of memory arrays 3601arranged in rows and columns of memory arrays 3601. Each memory array3601 comprises a plurality of memory subarrays 3602, a plurality oflocal sense amplifiers 3604, and a plurality of global sense amplifiers3606. In one embodiment, the local sense amplifier 3604 is disposedadjacent to a memory subarray 3602. In another embodiment, the localsense amplifier 3604 is shared between a plurality of memory subarrays3602. The local sense amplifier 3604 reads the contents of the memorycells within the corresponding memory subarray 3602. The memorysubarrays 3602 are arranged in rows and columns. The local senseamplifiers 3604 coupled to a column of memory subarrays 3602 are coupledto a global sense amplifier 3606. The memory cells may include redundantcells, reference cells, or spare cells. The local sense amplifier 3604may include an autozero function. In one embodiment, the global senseamplifiers 3606 include an autozero function. In another embodiment, theglobal sense amplifiers 3606 does not include an autozero function. Inanother embodiment, the memory system 3600 includes only the globalsense amplifiers 3606 at the top level (at the system 3600), which areshared between the memory arrays 3601.

FIG. 36A is a block diagram illustrating a memory system 3650 for amultilevel memory including local autozero sense amplifiers.

The memory system 3650 is similar to memory system 3600, but it does notinclude the global sense amplifiers 3606. In this case, memory subarrays3651 are similar to the memory subarrays 3601 without the global senseamplifiers 3606, but include buffers 3652 that are disposed locallyright after the local sense amplifier 3604 to drive a global bus (notshown) coupled between a sensed and amplified output of the buffer 3652to global latches (not shown) or output buffers (not shown). Thisarchitecture may be most realizable for highest speed. Alternatively,the latches instead of global latches may be disposed locally next tothe local sense amplifier 3604.

The memory subarrays 3602 of FIGS. 36 and 36A may be segmented arrays.The memory subarray 3602 may include status cells disposed in a separaterow or rows or same row which indicates status of the subarray/row suchas it is used for data or code storage, whether the subarray/row is aterase or program state, whether the subarray/row is good, not-so-good orbad condition, a number of bad cells in a subarray/row, or degree ofcell storage level wearing, or operational status such as bias valuesfor erase/program/read bias for each row or page. The memory subarray3602 may include reference cells disposed in a separate row or rows,which are enabled when a data row is enabled in a verify or a read mode.

The memory subarray 3602 may include reference cells that are read foreach page, such as described above, or for each segment.

The local current sense amplifier 3604 may be a current sense amplifier4000 (FIG. 40), the current sense amplifier 4100 (FIG. 41), thetwo-stage current sense amplifier 4200 (FIG. 42), a two-stage currentsense amplifier 4300 (FIG. 43), a two-stage indirect current sensingamplifier 4400 (FIG. 44), and a two-stage indirect current sensingamplifier 4500 (FIG. 45).

FIG. 37 is a block diagram illustrating a memory system 3700 includingsingle ended autozero sense amplifiers.

A memory system 3700 comprises a plurality of segmented arrays 3702, aplurality of autozero local sense amplifiers 3704, and a plurality ofautozero global sense amplifiers 3706. The memory system 3700 may bearranged in a manner similar to the memory system 3600 described above.For clarity, FIG. 37 shows only one segmented array 3702, one autozerolocal sense amplifier 3704, and one autozero global sense amplifier3706. The segmented array 3702 comprises a plurality of data memorycells 3708 and a plurality of reference memory cells 3710. For clarityonly one data memory cell 3708 and one reference memory cell 3710 areshown. The data cells 3708 and the reference memory cells 3710 arecoupled to the corresponding autozero local sense amplifier 3704 forsensing the content of the data memory cell 3708 in comparison with thereference memory cell 3710. The autozero local sense amplifier 3704 maybe, for example, one of the sense amplifiers described below inconjunction with FIGS. 40 through 45. The autozero local sense amplifierand the autozero global sense amplifier are single ended amplifiers.

In another embodiment, the autozero local sense amplifier 3704 may be acurrent sensing autozero sense amplifier, and the autozero global senseamplifier 3706 may be a voltage sensing autozero sense amplifier, suchas described above. In another embodiment, the autozero local senseamplifier 3704 may be a current sensing autozero sense amplifier, andthe autozero global sense amplifier 3706 may be a current sensingautozero sense amplifier. In another embodiment, the autozero localsense amplifier 3704 may be a voltage sensing autozero sense amplifier,and the autozero global autozero sense amplifier 3706 may be a currentsensing autozero sense amplifier.

FIG. 38 is a block diagram illustrating a memory system 3800 includingdifferential autozero sense amplifiers.

The memory system 3800 comprises a plurality of segmented memory arrays3802, a plurality of local autozero sense amplifiers 3804, and aplurality of global sense amplifiers 3806. For clarity, FIG. 38 showsonly one segmented memory array 3802, one autozero local sense amplifier3804, and one autozero global sense amplifier 3806. The memory system3800 may be arranged in a manner similar to the memory systems 3600 and3700, except that the autozero local sense amplifier 3804 and theautozero global sense amplifier 3806 may include a differential autozerosense amplifier such as the global sense amplifier 3400 described abovein conjunction with FIG. 34. In another embodiment, the memory system3800 does not include a global sense amplifier 3806. In this case, adifferential to single ended output and buffered stage may be coupledlocally right after the local sense amplifier 3804 to drive a global bus(not shown) coupled between from a sensed and amplified output of thebuffered stage to global latches (not shown) or output buffers (notshown).

FIG. 39 is a block diagram illustrating a memory system 3900 includingcrossed bitlines.

The memory system 3900 comprises a plurality of memory arrays 3902, anda plurality of global sense amplifiers 3904. For clarity and simplicity,only one memory array 3902 and one global sense amplifier 3904 areshown. The memory 3902 comprises a plurality of data cells 3906, aplurality of reference cells 3908 and a plurality of local senseamplifiers 3910. For clarity and simplicity, only one column of datacells 3906, one column of reference cells 3908, and one local senseamplifier 3910 are shown. A data bitline 3912 couples a column of datacells 3906 to the local sense amplifier 3910. A reference bitline 3914couples a column of reference cells 3908 to the local sense amplifier3910. The local sense amplifier 3910 is coupled to the global senseamplifier 3904. The data bitline 3912 and the reference bitline 3914 aredisposed in a crossed configuration with the bitlines 3912 and 3914being disposed so the signal path goes back and forth between thephysical location of the columns of data cells 3906 and reference cells3908. Crossing may provide similar electrical characteristics as atwisted wire pair. A global bitline 3916 couples the global senseamplifier 3904 to the local sense amplifier 3910. Global bitlines 3916may be arranged in a crossed configuration. One bitline 3912 or 3914 maybe crossed in the same metal while the other bitline makes a crossingjump by another metal line, a poly line, or a diffusion over the firstbitline.

In another embodiment, the local sense amplifiers 3910 and the globalsense amplifiers 3904 may use the sense amplifiers described below inconjunction with FIGS. 40–48.

FIG. 40 is a block diagram illustrating a current sense amplifier 4000including auto-zero.

The current sense amplifier 4000 uses autozero or equalization toequalize voltages on an output terminal 4099 and a node 4098 that storesa voltage corresponding to the current on a reference cell bitline (IR)4005 so that the signal path through the current sense amplifier 4000 issimilar for both the data cell bitline (ICELL) 4006 and the referencecell bitline (IR) 4005. The current sense amplifier 4000 may be used inthe autozero local sense amplifier 3704 of FIG. 37.

The current sense amplifier 4000 comprises a plurality of inverters 4001and 4002, a plurality of PMOS transistors 4010 and 4011, a plurality ofNMOS transistors 4020 through 4023, and a charge cancellation injectioncircuit 4036.

The NMOS transistors 4021 and 4023 are arranged as a switch toselectively couple the reference cell bitline (IR) 4005 or the data cellbitline (ICELL) 4006 to a first node 4007. The drain-source terminals ofthe NMOS transistor 4021 couple the reference cell bitline (IR) 4005 tothe first node 4007 in response to a first autozero selection signal4003 applied to a gate of the NMOS transistor 4021. The drain-powerterminals of the NMOS transistor 4023 couple the data cell bitline(ICELL) 4006 to the first node 4007 in response to a second autozeroselection signal 4004 applied to a gate of the NMOS transistor 4023. Inanother embodiment, PMOS transistors (not shown) may be coupled inparallel to the NMOS transistors 4021 and 4023 and controlled by thesecond autozero selection signal 4004 and the first autozero selectionsignal 4003, respectively.

The inverters 4001 and 4002 are coupled in series to generate the secondautozero selection signal 4004 and the first autozero selection signal4003, respectively, in response to an autozero control signal 4008applied to the input of the inverter 4001. In one embodiment, the timingof the generation of the first and second autoselection signals 4003 and4004, respectively, causes the NMOS transistors 4021 and 4023 operatingas switches to break before make.

The drain-source terminals of the diode connected PMOS transistor 4011couple a supply voltage VSUP to the first node 4007 to generate acurrent indicative of the voltage on the first node 4007 andcorrespondingly indicative of the voltage in the respective data memorycell or reference memory cell.

The drain-source terminals of the PMOS transistor 4010 and the NMOStransistor 4020 are coupled in series between the supply voltage andground to form an output terminal 4099 formed of the common node of thedrains of the transistors 4010 and 4020. The gate of the PMOS transistor4010 is coupled to the common node of the gate and drain of the PMOStransistor 4011 to form a current mirror with the PMOS transistor 4011.

The NMOS transistor 4022 is arranged as a switch to selectively couplethe voltage on the common node formed of the output terminal 4099 andthe drain terminals of the transistors 4010 and 4020 to a second node4098 in response to the first autozero selection signal 4003. When thevoltage on the output terminal 4099 is coupled to the second node 4098,the voltage on the second node 4098 is indicative of the voltage on thefirst node 4007, which corresponds to the data cell plus any offsetthrough the data path and the sense amplifier 4000. The drain-sourceterminals of the NMOS transistor 4022 couple the drain of the NMOStransistor 4020 to the common node formed of the second node 4098 andthe gate of the NMOS transistor 4020 to diode connect the NMOStransistor 4020 in response to the first autozero selection signal 4003applied to the gate of the NMOS transistor 4022.

In another embodiment, a capacitor (not shown) is coupled between thesecond node 4098 and ground to store charge indicative of the referencememory cell current.

The charge injection cancellation circuit 4036 provides charge injectioncancellation caused by the NMOS transistor 4022 switching. The chargeinjection cancellation circuit 4036 may be an NMOS transistor arrangedas a capacitor coupled between the source of the NMOS transistor 4022and the second autozero selection signal 4004. In one embodiment, theNMOS transistor 4036 is one-half the size of the NMOS transistor 4022.In another embodiment, the drain-source terminals of a PMOS transistor(not shown) may be coupled between the drain-source terminals of theNMOS transistor 4022.

As an overview, the NMOS transistors 4021 and 4023 selectively couplethe reference memory cell line (IR) 4005 and the data memory cell(Icell) 4006 to the first node 4007 in response to the first and secondautozero selection signals 4003 and 4004, respectively. The data fromthe reference memory cell line 4005 and the data memory cell line 4006may be mismatched, but after the data reaches the first node 4007, thecurrent sense amplifier 4000 provides a similar path to eliminatemismatch of the signals from the data memory cell and the referencememory cell when they are compared. The reference level of the referencememory cell is first converted from a current to a voltage and acorresponding voltage is stored on the second data node 4098 and thenthe data cell is read by applying the current to the first node 4007 andcomparing to the reference memory cell stored on the second node 4098 toproduce an output signal on the output terminal 4099 indicative of thedifference in the voltage on the first node 4007 and the second node4098 to indicate the difference between the data memory cell and thereference voltage from the reference memory cell.

During a first operation, the current sense amplifier 4000 is operatedto store a voltage on the second node 4098 indicative of the referencebitline (IR) 4005. The first autozero selection signal 4003 is set to anenable state, and the second autozero selection signal 4004 iscorrespondingly set to the disabled state. When the first autozeroselection signal 4003 is enabled, the NMOS transistors 4021 and 4022 areturned on. The NMOS transistor 4021 applies the reference current (IR)to the first node 4007 which is applied to the PMOS transistor 4011. ThePMOS transistor 4010 mirrors the current from the PMOS transistor 4011.The NMOS transistor 4020 is diode connected because the enabled NMOStransistor 4022 shorts the output terminal to the second node 4098. Thevoltage on the second node 4098 sustains the current in the PMOStransistor 4010 and the NMOS transistor 4020. The voltage on the secondnode 4098 corresponds to the reference memory cell current IR plus anyoffset between the PMOS transistors 4011 and 4010, hence the current inthe PMOS transistor 4010 corresponds to the reference memory cellcurrent IR but not necessarily exactly due to any mismatch between thePMOS transistors 4010 and 4011.

During the second operation, the first autozero selection signal 4003 isdisabled and the second autozero selection signal 4004 is enabled, toconnect the data cell to the first node 4007. The NMOS transistor 4023is turned on and the NMOS transistors 4021 and 4022 are turned off. TheNMOS transistor 4023 provides the data cell current ICELL to the PMOStransistor 4011, which the PMOS transistor 4010 mirrors. The PMOStransistor 4010 compares this current to the current from the NMOStransistor 4020 generated in response to the voltage on the second node4098. The output voltage VOUT on the output terminal 4099 is the currentdifference between the two currents multiplied by the output impedancelooking into the PMOS transistor 4010 and the NMOS transistor 4020. Inbiased voltage range, the output impedance is the drain-sourceresistance of the PMOS transistor 4010 in parallel with the drain-sourceresistance of the NMOS transistor 4020.

FIG. 41 is a block diagram illustrating a current sense amplifier 4100including autozero and replica loading.

The current sense amplifier 4100 is similar to the current senseamplifier 4000, and includes a replica loading circuit comprising a PMOStransistor 4114 and an NMOS transistor 4127 that are arranged toprecharge the data cell reference line (ICELL) 4006. Like numbersrepresent like elements. The current sense amplifier 4100 may be used inthe autozero sense amplifier 3704 of FIG. 37.

The drain-source terminals of the diode connected PMOS transistor 4114and the NMOS transistor 4127 are coupled in series between the supplyvoltage Vsup and the data memory cell line (ICELL) 4006. The NMOStransistor 4127 is enabled by the first autozero selection signal 4003applied to the gate thereof. The transistors 4114 and 4127 replicate theloading of the transistors 4011 and 4021. In one embodiment, the PMOStransistor 4114 is the same size as the PMOS transistor 4011. Likewise,the NMOS transistor 4127 is the same size as the NMOS transistor 4021.

When the first autozero selection signal 4003 is enabled, both the NMOStransistors 4021 and 4127 are enabled. The first node 4007 is brought tothe level of the reference memory cell line (IR) 4005 as describedabove, and the data cell line (ICELL) 4006 is precharged. After thesecond autozero selection signal 4004 is enabled, the NMOS transistor4127 is disabled and the first node 4007 is brought to the data cellreference as described above, but at a faster speed because of theprecharge.

FIG. 42 is a block diagram illustrating a two stage current senseamplifier 4200 including autozero.

The two-stage current sense amplifier 4200 is similar to the currentsense amplifier 4100 of FIG. 41, but also includes an output stage. Theoutput stage is autozeroed or equalized to the output of the currentsense amplifier 4100 during a first operation. The two-stage currentsense amplifier 4200 may be used in the autozero sense amplifier 3704 ofFIG. 37.

The two stage current sense amplifier 4200 comprises a current senseamplifier 4100 and an output stage 4202.

The output stage 4202 compares or amplifies the output of the currentsense amplifier 4100. The output stage 4202 comprises a PMOS transistor4215, NMOS transistors 4228 and 4229, and a charge injectioncancellation circuit 4237. The drain-source terminals of the PMOStransistor 4215 and the NMOS transistor 4229 are coupled in seriesbetween the supply voltage and ground, and form an output terminal 4299at a common node formed of the drains of the transistors 4215 and 4229.The gate of the PMOS transistor 4215 is coupled to the gate of the PMOStransistor 4010 for biasing that is the same as the same autozero biascondition. The gate of the NMOS transistor 4229 is biased by the output4199 of the current sense amplifier 4100. The drain-source terminals ofthe NMOS transistor 4228 diode connect the NMOS transistor 4229 inresponse to the first autozero selection signal 4003. In one embodiment,the transistors 4215, 4229, and 4228 are similar to respectivetransistors 4010, 4020, and 4022 to increase gain and speed. The chargeinjection cancellation circuit 4237 may be an NMOS transistor arrangedas a capacitor coupled between the source of the NMOS transistor 4228and the second autozero selection signal 4004. In one embodiment, theNMOS transistor of the charge injection cancellation circuit 4237 issimilar to the charge injection cancellation circuit 4036.

In an alternative embodiment, a decoupling capacitor (not shown) may becoupled between the output of the current sense amplifier 4100 and thecommon node formed of the gate of the NMOS transistor 4229, the sourceof the NMOS transistor 4228 and the charge injection cancellationcircuit 4237.

FIG. 43 is a block diagram illustrating a two-stage current senseamplifier 4300 including autozero.

The two-stage current sense amplifier 4300 is similar to the currentsense amplifier 4000 of FIG. 40, but also includes an output stage. Theoutput stage is autozeroed or equalized to the output of the currentsense amplifier 4000 during a first operation. The two-stage currentsense amplifier 4300 may be used in the autozero local sense amplifier3704 of FIG. 37.

The two-stage current sense amplifier 4300 comprises the current senseamplifier 4000 and an output stage 4302.

The output stage 4302 inverts and amplifies the output signal from thecurrent sense amplifier 4000. The output stage 4302 comprises a PMOStransistor 4315, NMOS transistors 4328 and 4329, a charge injectioncancellation circuit 4337, and a capacitor 4340. The drain-sourceterminals of the PMOS transistor 4315 and the NMOS transistor 4329 arecoupled in series between the supply voltage and ground, and form anoutput terminal 4399 at the common node formed of the drains of thetransistors 4315 and 4329. The gates of the transistors 4315 and 4329are coupled together to form an inverter of the transistors 4315 and4329. The drain-source terminals of the NMOS transistor 4328 diodeconnect the NMOS transistor 4329 in response to the first autozeroselection signal 4003. The charge injection cancellation circuit 4337may be an NMOS transistor arranged as a capacitor coupled between thesource of the NMOS transistor 4328 and the second autozero selectionsignal 4004. The capacitor 4340 is coupled between the output 4099 ofthe current sense amplifier 4000 and the common node formed of the gatesof the transistors 4315 and 4329, the source of the NMOS transistor 4328and the charge injection cancellation circuit 4337. The capacitor 4340decouples the output stage 4302 from the current sense amplifier 4000.

In an alternative embodiment, the gate of the PMOS transistor 4315 maybe coupled to the PMOS transistor 4010.

FIG. 44 is a block diagram illustrating a two-stage indirect currentsensing amplifier 4400 having autozero.

The two-stage indirect current sensing amplifier 4400 may be used in theautozero local sense amplifier 3704 of FIG. 37.

The two-stage indirect current sensing amplifier 4400 comprises anindirect current input stage 4401, an indirect current sense amplifier4402, and an output stage 4403. The indirect current input stage 4401selectively switches between a reference memory cell bitline (IREF) 4495and a data memory cell bitline (ICELL) 4496. In a first operation, theindirect current input stage 4401 stores a voltage corresponding to thecurrent on the reference memory cell bitline (IREF) 4495 and any offsetin the circuit. The two-stage indirect current sensing amplifier 4400autozeroes or equalizes the output of the indirect current senseamplifier 4402 and the output of the output stage 4403 with the storedvoltage. During a second operation, the indirect current input stage4401 couples the data memory cell bitline (ICELL) 4496 to an input ofthe indirect current sense amplifier 4402 for comparison with thereference current on the reference memory cell bitline (IREF) 4495 asadjusted by the stored voltage in the indirect current input stage 4401.The indirect current sense amplifier 4402 amplifies the voltagedifference, which is further amplified by the output stage 4403.

The indirect current input stage 4401 comprises a plurality of NMOStransistors 4421 through 4424 and a capacitor 4433. The indirect currentsense amplifier 4402 comprises a plurality of PMOS transistors 4410 and4419, a plurality of NMOS transistors 4427 through 4429, and a chargeinjection cancellation circuit 4435. The output stage 4403 comprises aPMOS transistor 4411, a plurality of NMOS transistors 4420 and 4426, anda charge injection cancellation circuit 4436.

The diode connected NMOS transistor 4421 couples the reference memorycell bitline (IREF) 4495 to ground. The reference memory cell bitline(IREF) 4495 is coupled to the gate of the NMOS transistor 4428 forproviding a reference bias and also is coupled to the drain of the NMOStransistor 4423 for selective switching to the capacitor 4433 inresponse to a first autozero selection signal 4493.

The diode connected NMOS transistor 4422 couples the data memory cellbitline (ICELL) 4496 to ground. The data memory cell bitline (ICELL)4496 is coupled to the drain of the NMOS transistor 4424 for selectiveswitching to the capacitor 4433 in response to a second autozeroselection signal 4494.

The drain-source terminals of the diode connected PMOS transistor 4419and the NMOS transistor 4428 are coupled in series between the supplyvoltage VSUP and ground to provide a reference current in response tothe reference bias applied to the gate of the NMOS transistor 4428 bythe reference memory cell bitline (IREF) 4495. The drain-sourceterminals of the PMOS transistor 4410 and the NMOS transistor 4429 arecoupled in series between the supply voltage VSUP and ground. The gateof the PMOS transistor 4410 is coupled to the common node formed of thegate and drain of the PMOS transistor 4419 to form a current mirror withthe PMOS transistor 4419. The drain-source terminals of the NMOStransistor 4427 diode connect the NMOS transistor 4429 in response tobeing enabled by the first autozero selection signal 4493. The chargeinjection cancellation circuit 4435 is coupled to the source of the NMOStransistor 4427 to provide charge injection cancellation in response tothe second autozero selection signal 4494. The charge injectioncancellation circuit 4435 may be an NMOS transistor arranged as acapacitor coupled between the source of the NMOS transistor 4427 and thesecond autozero selection signal 4494.

In the output stage 4403, the drain-source terminals of the PMOStransistor 4411 and the NMOS transistor 4420 are coupled in seriesbetween the supply voltage VSUP and ground, and the drains of thetransistors 4411 and 4420 form an output terminal 4499. The gate of thePMOS transistor 4411 is coupled to the common node formed of the gateand drain of the PMOS transistor 4419 to form a current mirror with thePMOS transistor 4419. The drain-source terminals of the NMOS transistor4426 diode connect the NMOS transistor 4420 and couple the output 4459of the indirect current sense amplifier 4402 to the output terminal 4499of the output stage 4403 in response to being enabled by the firstautozero selection signal 4493. The charge injection cancellationcircuit 4436 is coupled to the source of the NMOS transistor 4426 toprovide charge injection cancellation in response to the second autozeroselection signal 4494. The charge injection cancellation circuit 4436may be an NMOS transistor arranged as a capacitor coupled between thesource of the NMOS transistor 4426 and the second autozero selectionsignal 4494.

During the first operation, the first autozero selection signal 4493 isenabled, and the transistors 4423, 4427, and 4426 are enabled to couplethe reference memory cell bitline (IREF) 4495 to the capacitor 4433which stores the voltage corresponding to the current on the referencememory cell bitline (IREF) 4495 and any offset in the circuit, andcouples the voltage to the output of the indirect current senseamplifier 4402 and the output 4499 of the output stage 4403. During asecond operation, the second autozero selection signal 4494 is enabled,which enables the NMOS transistor 4424 to couple the data memory cellbitline (ICELL) 4496 to the capacitor 4433, which is compared to thestored voltage. The indirect current sense amplifier 4402 amplifies thevoltage difference, which is further amplified by the output stage 4403.

The mismatch between the NMOS transistors 4421 and 4422 may not becancelled in the two-stage indirect current sense amplifier 4400.

FIG. 45 is a block diagram illustrating a two-stage indirect currentsensing amplifier 4500 having autozero.

The two-stage indirect current sensing amplifier 4500 is similar to theindirect current sensing amplifier 4400, but instead includes aninverter arranged output stage. The output stage is autozeroed orequalized to the output of an indirect current sense amplifier during afirst operation. The two-stage indirect current sensing amplifier 4500may be used in the autozero local sense amplifier 3704 of FIG. 37.

The two-stage indirect current sense amplifier 4500 comprises anindirect current input stage 4401, an indirect current sense amplifier4402 and an output stage 4503. The output stage 4503 comprises a PMOStransistor 4511, a plurality of NMOS transistors 4520 and 4526, acapacitor 4532, and a charge injection cancellation circuit 4536. Thetransistors 4511 and 4520 are arranged as an inverter gain stage withself bias. The drain-source terminals of the PMOS transistor 4511 andthe NMOS transistor 4520 are coupled in series between the supplyvoltage VSUP and ground, and include gates coupled to each other. Thedrains of the transistors 4511 and 4520 form an output node 4599. Thecapacitor 4532 couples the output of the indirect current senseamplifier 4402 to the common node formed of the gates of the transistors4511 and 4520 to decouple the indirect current sense amplifier 4402 andthe output stage 4503. The drain-source terminals of the NMOS transistor4526 couple the output terminal of the output stage 4503 to the commonnode formed of the gates of the transistors 4511 and 4520 in response tothe first autozero selection signal 4493. The charge injectioncancellation circuit 4536 is coupled to the source of the NMOStransistor 4526 in response to the second autozero selection signal4494. The charge injection cancellation circuit 4536 may be an NMOStransistor arranged as a capacitor coupled between the source of theNMOS transistor 4526 and the second autozero selection signal 4494.

The memory system 3700 of FIG. 37 may be configured to operate at lowvoltages, e.g. 0. to 1.2 volts. The local sense amplifier 3706 mayinclude a readout circuit that operates to read multilevel memory cellsin this voltage range, such as described below in conjunction with FIGS.46–48.

FIG. 46 is a block diagram illustrating a memory system 4600 including alow voltage sense amplifier. The sensing shown in FIGS. 46, 46A, 47,47A, 47B, 48, 48A, and 48B is to be known as Direct (Memory) CellSensing because a sensing element (the load) is connected directlythrough decoding circuitry to the memory cell but not through bias(cascading transistors. The sensing elements are capable of sub-volt(less than 1 volt power supply) sensing. In one embodiment, the circuitsof FIGS. 46, 46A, 47, 47A, 47B, 48, 48A, and 48B are coupled to acomparison circuit that uses capacitors for autozero, signal coupling,and signal comparison, such as shown in FIGS. 34, 35, 38, and 57.

The memory system 4600 is similar to the memory system 3600 describedabove in conjunction with FIG. 36, but the local sense amplifier 3604includes a readout circuit 4602. For clarity, FIG. 46 shows only onememory subarray 3602 and one local sense amplifier 3604, and only onememory cell 4603 is shown in the memory subarray 3602. The readoutcircuit 4602 may operate in a low voltage range. The readout circuit4602 may read memory cells that store low voltages and may provide aread signal 4604 indicative of the content of the memory cells 4603.

The readout circuit 4602 comprises a buffer 4606 and a resistor 4608.The resistor 4608 provides feedback between an output and an invertinginput of the buffer 4606. The inverting input of the buffer 4606 iscoupled to the bitline for coupling to the memory cells 4603. Anon-inverting input of the buffer 4606 is coupled to a reference voltagefrom a reference memory cell (not shown).

As an illustrative example, the voltage stored in the memory cell 4603and the voltage (VBITLN) on the bitline may be in the range of 0.0through 1.0 volts. The minimum supply voltage (VDD_(min)) equals thevoltage (VBITLN) on the bitline plus a differential operating voltage(dVOP), for example 0.5 volts. A control gate voltage (VCGR) of 1.8 to2.4 volts is applied to the control gate of the memory cell 4603. Thememory cell 4603 operates in a linear region or saturation. During aread or verify, the bitline voltage (VBITLN) may be 0.2 V or 0.6 V. Theread signal 4604 output from the comparator 4606 may be in a range of0.2 to 0.4 volts or 0.8 to 1.2 volts.

FIG. 46A is a block diagram illustrating a memory system 4650 includinga low voltage sense amplifier.

The memory system 4650 is similar to the memory system 4600 describedabove in conjunction with FIG. 46, but the local sense amplifier 3604includes a readout circuit 4652 that may operate in a low voltage range.The readout circuit 4602 may read memory cells that store low voltagesand may provide a read signal 4654 indicative of the content of thememory cells 4603. The readout circuit 4652 comprises a resistor 4608coupled between the supply voltage and the output 4654, which is coupledto the bitline coupled to the memory cell 4603.

FIG. 47 is a block diagram illustrating a memory system 4700 including alow voltage sense amplifier.

The memory system 4700 is similar to the memory system 4600 in FIG. 46described above. For clarity, FIG. 47 shows only one memory subarray3602 and one local sense amplifier 3604. The local sense amplifier 3604includes a readout circuit 4702, which is similar to the readout circuit4602, except that a PMOS transistor 4708 functions as the feedbackelement and replaces the resistor 4608. The readout circuit 4702 mayoperate in a low voltage range. The PMOS transistor 4708 includesdrain-source terminals coupled between the output and the invertinginput of a buffer 4706 and includes a gate coupled to a fixed voltage,such as ground.

FIG. 47A is a block diagram illustrating a memory system 4750 includinga low voltage sense amplifier.

The memory system 4750 is similar to the memory system 4700 in FIG. 47described above, but the local sense amplifier 3604 includes a readoutcircuit 4752 that comprises a PMOS transistor 4758 that includesdrain-source terminals coupled between a supply voltage and an outputnode 4754, which is coupled to the bitline coupled to the memory cell4603, and includes a gate coupled to a fixed voltage, such as ground.

FIG. 47B is a block diagram illustrating a memory system 4770 includinga low voltage sense amplifier.

The memory system 4770 is similar to the memory system 4750 in FIG. 47Adescribed above, but the local sense amplifier 3604 includes a readoutcircuit 4772 that comprises a diode connected PMOS transistor 4758coupled between the supply voltage and an output node 4774.

FIG. 48 is a block diagram illustrating a memory system 4800 including alow voltage sense amplifier.

The memory system 4800 is similar to the memory system 4700 of FIG. 47.For clarity, FIG. 48 shows only one memory subarray 4602 and one localsense amplifier 3604. The local sense amplifier 3604 includes a readoutcircuit 4802 that comprises a buffer 4806 and an NMOS transistor 4808,which can be an enhancement NMOS transistor (threshold voltage VT=˜0.5to 1.0V) or a native NMOS transistor (threshold voltage VT=−0.2 to 0.2).The NMOS transistor 4808 includes drain-source terminals coupled betweenthe supply voltage VDD and an inverting input of the buffer 4806, andincludes a gate coupled to the output of the buffer 4806 for feedback.The operation of the readout circuit 4802 is similar to that describedabove for the readout circuit 4602, except the minimum supply voltageVDD_(min) equals a fixed voltage, e.g., 0.4 volts, plus a differentialoperating voltage (dVOP), e.g., 0.5 volts.

FIG. 48A is a block diagram illustrating a memory system 4850 includinga low voltage sense amplifier.

The memory system 4850 is similar to the memory system 4800 in FIG. 48described above, but the local sense amplifier 3604 includes a readoutcircuit 4852 that comprises a NMOS transistor 4858 that includesdrain-source terminals coupled between a supply voltage and an outputnode 4854, which is coupled to the bitline coupled to the memory cell4603, and includes a gate coupled to a fixed voltage.

FIG. 48B is a block diagram illustrating a memory system 4870 includinga low voltage sense amplifier.

The memory system 4870 is similar to the memory system 4850 in FIG. 48Adescribed above, but the local sense amplifier 3604 includes a readoutcircuit 4872 that comprises a diode connected NMOS transistor 4878coupled between the supply voltage and an output node 4874.

FIG. 49 is a schematic diagram illustrating a shared sense amplifiersegmented reference array 4900. The shared sense amplifier segmentedreference array 4900 may be used in the memory system 3800 describedabove in conjunction with FIG. 38.

The shared sense amplifier segmented reference array 4900 comprises aplurality of array segments 4902-1 and 4902-2 and a plurality ofdifferential sense amplifiers 4904-1 through 4904-3. As an illustrativeexample, the array 4900 includes two array segments 4902. In oneembodiment, the array segments 4902 are disposed above and below thedifferential sense amplifiers 4904.

The array segment 4902 comprises a plurality of memory cells 4912arranged in rows and columns. A pair of reference rows 4913 comprise tworows of memory cells that store reference levels. Word lines WLR0 andWLR1 are coupled to even and odd reference rows, respectively, toaccount for odd and even row effect. Even or odd reference rows are usedfor even or odd data rows respectively. Word lines WL0 through WL3 arecoupled to the data rows.

The array segment 4902 comprises a plurality of memory columns 4906-0through 4906-7, a first multiplexer 4908 and a plurality of secondmultiplexers 4910-1 through 4910-3. As an illustrative example, thememory cells store two bits, and accordingly, there are three referencecells for three reference levels. In this case, the memory cells 4912 ofthe reference rows 4913 store reference voltages in the memory cells inthe three memory columns 4906-5-through 4906-7. The other memory cells4912 in the reference row 4913 may be left floating, or may beconnected, but not used. The first multiplexer 4908 couples each of thememory columns 4906 to a first input of each of the differential senseamplifiers 4904. The multiplexers 4910-1 through 4910-3 couple therespective memory column 4906-5 through 4906-7 to a second input of arespective differential sense amplifier 4904-1 through 4904-3. Eachmemory column 4906 comprises a column of the memory cells 4912 and amultiplexer 4914. For clarity, reference numerals are shown only for onememory column 4906. The multiplexer 4914 allows the memory cells 4912 tobe accessed.

In one embodiment, data is stored in one of the array segments 4902, forexample, the bottom array segment 4902, and reference voltages arestored in the memory cells 4912 of the reference rows 4913 that are inthe memory columns 4906-5 through 4906-7 of the other array segment4902, for example, the top array segment 4902. When memory cells 4912 inone of the array segments 4902 are selected for multiplexing to thedifferential sense amplifiers 4904, memory cells 4912 in the memorycolumns 4906-5 through 4906-7 that function as reference memory cells ofthe other array segment 4902 is selected at the same time.

FIG. 50 is a schematic diagram illustrating a memory cell replica senseamplifier 5000.

The memory cell replica sense amplifier 5000 comprises a plurality ofmemory cell circuits 5002-0 and 5002-1, a plurality of replica memorycell circuits 5004-0 through 5004-4, a differential amplifier 5006, anda plurality of bias generators 5008-0 and 5008-1. For clarity, thedetails of only the bias generator 5008-0 are shown. Although fivereplica memory cell circuits 5004 are shown, other numbers of circuits5004 may be used. One of inputs to the differential amplifier 5006couples to the read out voltage from a data memory cell and the otherinput couples to read out voltage from a reference memory cell.

The memory cell circuits 5002 comprise a memory cell circuit 5010, aPMOS transistor 5012, and a plurality of NMOS transistors 5014 and 5016.For clarity, the details of only the memory cell circuit 5002-0 areshown. The memory cell circuit 5010 is a circuit that is an equivalentmodel for a source side injection (SSI) split gate flash memory cell inread mode. In one embodiment, the memory cell circuit 5010 comprises apair of NMOS transistors, in which a bottom transistor corresponds to afloating gate transistor, and a top transistor corresponds to a selectgate (control gate) transistor. The NMOS transistor 5016 operates as aswitch or multiplexer to allow access to the memory cell circuit 5010 inresponse to a column decode (COLDEC) signal. The NMOS transistor 5014provides column bias to the memory cell circuit 5002 in response to thebias generator 5008, which generates a bias and includes feedbackcontrol of the bias.

The replica memory cell circuits 5004 comprise a replica memory cellcircuit 5020, a PMOS transistor 5022, and a plurality of NMOStransistors 5024 and 5026. For clarity, the details of only the replicamemory cell circuit 5004-0 are shown. The replica memory circuit 5020replicates the SSI flash memory cells. In one embodiment, the replicamemory cell circuit 5020 comprises a pair of NMOS transistors. The PMOStransistor 5022 mirrors the current of the PMOS transistor 5012 of thememory cell circuit 5002. The NMOS transistor 5026 replicates the columnselect decoding of the NMOS transistor 5016. The NMOS transistor 5024provides column bias to the replica memory cell circuit 5004. A controlgate voltage CG is applied to the memory cell circuits 5010 of bothmemory cell circuits 5002-0 and 5002-1. A data floating gate voltage FCDis shown as a storage node of the data memory cell 5010 of the memorycell circuit 5002-0. A reference floating gate voltage FCR is shown as astorage node of the reference memory cell 5010 of the memory cellcircuit 5002-1. The gate of a transistor in the replica memory cellcircuit 5020 is coupled to the drain of the NMOS transistor 5024 so thatthe output voltage is approximately the same as the floating gatevoltage because the size and operating condition of the replica memorycell circuit 5020 is equivalent to that of the memory cell circuit 5002.The drain of the PMOS transistor 5022 of the replica memory cell circuit5004-0 provides an output data voltage, and the drain of the PMOStransistor 5022 of the other replica memory circuits 5004 provides anoutput reference voltage. The PMOS transistors 5022 of the replicamemory cell circuits 5004-1 through 5004-4 are dimensioned to a ratio tothe PMOS transistor 5012 of the memory cell circuit 5002 to setdifferent output reference voltage levels. As an illustrative example,three levels are set for two-bit cells, and a fourth level is set as anerase reference.

In one embodiment, the transistors 5014 of the memory cell circuits 5002and the transistors 5024 of the replica memory cell circuits 5004 arethe same size.

In another embodiment, the replica memory cell circuits 5004-1 through5004-4 are coupled to a corresponding memory cell circuit 5002. In thiscase each memory cell circuit 5010 of the memory circuit 5002 has adifferent floating gate voltage FGR to generate different levels.

FIG. 51 is a schematic diagram illustrating a different current senseamplifier 5100.

The differential current sense amplifier 5100 comprises a plurality ofcurrent sources 5101 through 5106, a plurality of PMOS transistors 5108and 5109, and a plurality of NMOS transistors 5112 and 5113. The currentsense amplifiers 5101 and 5104 are arranged as a differential senseamplifier to form an up output (OP) node 5120 between the currentsources 5101 and 5103 and to form a down output (ON) node 5121 betweenthe current sources 5102 and 5104. The current source 5105 is parallelto the current source 5104, and the current source 5106 is parallel withthe current source 5103. The current source 5106 is a data currentsource. The current source 5105 is a reference current source. In oneembodiment, the current source 5105 replicates the reference currentIREF from a reference memory cell, and the current source 5106replicates the data current IDAT from a data memory cell.

The drain-source terminals of the PMOS transistor 5108 and the diodeconnected NMOS transistor 5112 are coupled between the up output node5120 and ground. The drain-source terminals of the PMOS transistor 5109and the NMOS transistor 5103 are coupled between the down output node5121 and ground. The drains of the transistors 5109 and 5113 form anoutput node 5136. A bias voltage VPBIAS applied to the gates of the PMOStransistors 5108 and 5109 establishes a bias point on the up node 5120and the down node 5121. The drain of the NMOS transistor 5112 biases thegate of the NMOS transistor 5113 to mirror the current.

In one embodiment, the fixed bias currents of the current sources 5101and 5102 are set equal (I₅₁₀₁=I₅₁₀₂). The fixed bias currents of thecurrent sources 5103 and 5104 are set equal to each other (I₅₁₀₃=I₅₁₀₄).The current of the current source 5101 is greater than the current ofthe current source 5103 (I₅₁₀₁>I₅₁₀₃). As an illustrative example, thecurrent source 5103 provides a current of 30 μa and the current source5101 provides a bias fixed current of 60 μa. The current in the NMOStransistor 5113, the current in the NMOS transistor 5112 and the currentin the PMOS transistors 5108 are equal to each other(I₅₁₁₃=I₅₁₁₂=I₅₁₀₈), and equal the difference of the current source 5101and the sum of the currents from the current sources 5103 and 5106(I₅₁₁₃=I₅₁₁₂=I₅₁₀₈=I₅₁₀₁−I₅₁₀₃−IDAT). This relationship follows from theNMOS transistor 5113 mirroring the current of the NMOS transistor 5112.The current from the PMOS transistor 5109 is the difference between thecurrent from the current source 5102 and the sum of the currents of thecurrent sources 5104 and 5105 (I₅₁₀₉=I₅₁₀₂−I₅₁₀₄−IREF). Accordingly, theoutput voltage equalsVOUT=ΔI*R _(OUT)=(I ₅₁₁₃ −I ₅₁₀₉)*R _(OUT),where the resistance R_(OUT) is the equivalent resistance at the outputnode (VOUT) 5116. Equivalently the output voltage equalsVOUT=(IREF−DAT)*R _(OUT),which is the difference of the data and reference currents multiplied bythe output resistance.Alternatively, the output from the output terminal 5136 may be a currentthat is the difference of the data and reference currents multiplied bya gain factor G, orIOUT=G(IDAT−IREF),where the gain factor G may be provided by an additional current gaincircuit (not shown) having a gain G.

FIG. 52 is a schematic diagram illustrating a two-stage differentialcurrent sense amplifier 5200.

The two-stage differential current sense amplifier 5200 comprises thedifferential current sense amplifier 5100 and an output stage 5202. Theoutput stage 5202 comprises a plurality of PMOS transistors 5204 and5205 and a plurality of NMOS transistors 5206 and 5207. The output stage5202 operates as another gain stage. The output stage 5202 provides anoutput rail to rail at an output node VOUT.

In another embodiment, the differential current sense amplifier stage5100 includes an NMOS transistor 5113 that has a gate that is diodeconnected instead of being coupled to the NMOS transistor 5112.

The drain-source terminals of the diode connected PMOS transistor 5204and the NMOS transistor 5206 are coupled in series between the supplyvoltage and ground. The gate of the NMOS transistor 5206 is coupled tothe drain of the NMOS transistor 5112. The drain-source terminals of thePMOS transistor 5205 and the NMOS transistor 5207 are coupled in seriesbetween the supply voltage and ground and form an output voltageterminal 5216 of the common node of the drains of the transistors 5205and 5207. The gate of the PMOS transistor 5205 is coupled to the drainof the PMOS transistor 5204 to mirror the current of the PMOS transistor5204. The gate of the NMOS transistor 5207 is coupled to the drain ofthe NMOS transistor 5113.

FIG. 53 is a schematic diagram illustrating a current difference senseamplifier 5300.

The differential current sense amplifier 5300 comprises a referencecurrent source 5302 that provides a reference current IREF, a datacurrent source 5304 that provides a data current IDAT, and an outputcurrent source 5306 that provides an output current IOUT. The referencecurrent source 5302 and the data current source 5304 are coupled inseries between a power terminal and a ground terminal and form an outputnode 5308 that provides an output that is a current. The output currentsource 5306 is coupled between the output node 5308 and ground.

If the data current IDAT is greater than the reference current IREF, theoutput current IOUT equals zero. Otherwise, the output currentIOUT=IREF−IDAT.

In another embodiment, the reference current source 5302 and the outputcurrent source 5306 may be interchanged.

In another embodiment, the output of the sense amplifier 5300 may be anoutput voltage VOUT equals the difference of the data and referencecurrent multiplied by the output resistance, orVOUT=V(OUT1)=(IREF−IDAT)*ROUT,where ROUT is the equivalent resistance at the output node 5308.

In another embodiment, the output current IOUT may be referred to thepositive rail instead of ground, coupled between the power terminal andthe output node 5308, and such the output current IOUT=IDAT−IREF.

In another embodiment, another output current source may be coupledbetween the power terminal and the output node 5308 to generate apositive output current IOUTP(=IDAT−IREF) in addition to a negativeoutput current IOUTN(=IREF−IDAT) from the output current source 5306.

FIG. 54 is a schematic diagram illustrating a current difference senseamplifier 5400.

The current difference sense amplifier 5400 comprises a referencecurrent source 5402 that provides a reference current IREF, a datacurrent source 5404 that provides a data current IDAT, a PMOS transistor5406, and a plurality of NMOS transistors 5408 and 5410. The NMOStransistor 5410 provides an output current IOUT. The reference currentsource 5402 and the data current source 5404 are coupled in seriesbetween a power terminal and a ground terminal and form an output node5412. The drain-source terminals of the PMOS transistor 5406 and thediode connected NMOS transistor 5408 are coupled in series between theoutput node 5412 and ground. The drain-source terminals of the NMOStransistor are coupled between an output note 5414 and ground, and thegate of the NMOS transistor 5410 is biased by the drain of the PMOStransistor 5406, which has a bias voltage VPBIAS applied to the gate ofthe PMOS transistor 5406 to establish the bias voltage on the node 5412.In another embodiment, the diode connected NMOS transistor 5408 isconnected directly to the node 5412, i.e., without coupling through thePMOS transistor 5406.

The transistors 5406, 5408, and 5410 form an output stage to buffer thecurrent, and amplify a current difference. The current flow in the NMOStransistor 5408 equals the difference of the reference current IREF andthe data current IDAT, or I5408=IREF−IDAT.

The output current IOUT equals the difference of the data and referencecurrents multiplied by a gain factor G, orIOUT=G(IDAT−IREF).

The size of the NMOS transistor 5410 equals the gain factor G times thesize of the NMOS transistor 5408.

FIG. 55 is a schematic diagram illustrating a dynamic sense amplifier5500.

The dynamic sense amplifier 5500 comprises a data memory cell 5502, areference memory cell 5504, a plurality of NMOS transistors 5506, 5508,5510, 5512, 5514, 5516, and a comparator 5518. For clarity, only onedata memory cell 5502 and one reference memory cell 5504 are shown forsubarray, but a subarray comprises a plurality of data memory cells 5502arranged in columns and a plurality of reference cells 5504 arranged incolumns. Each column of data memory cells 5502 includes correspondingNMOS transistors 5506 and 5508 for decoding. Each reference columncomprises NMOS transistors 5510 and 5512 for decoding. Only one columnwith one data memory cell 5502 is shown.

The comparator 5518 determines the voltage of the data memory cell 5502by comparing the cell voltage (VCELLD) on a cell bitline 5520 to areference voltage (VCELLR) on a reference bitline 5522. The NMOStransistor 5514 couples a bias voltage (VBIAS) 5524 to the data cellvoltage 5520 in response to an initialize bitline (INITBL) signal 5526.The NMOS transistor 5516 couples the bias voltage signal 5524 to thereference cell voltage 5522 in response to the initialized bitlinesignal 5526. The cell bitline 5520 has a capacitance shown as acapacitor 5528. The reference bitline 5522 has a capacitance shown as acapacitor 5530. Additional capacitance can be added to the bit lines5520 and 5522 to achieve a desired value of capacitance. A data controlgate voltage (VCGR) 5532 is applied to the control gate of the datamemory cell 5502. A reference control voltage (VCGEFR) 5534 is appliedto the control gate of the reference memory cell 5504. A comparatorenable (ENBLADIFA) signal 5536 enables the comparator 5518.

FIG. 56 is a graph illustrating the control signals and voltages of thedynamic sense amplifier 5500.

The initialize bitline signal 5526 is set to high to enable the NMOStransistors 5514 and 5516 to initialize the bitlines 5520 and 5522,respectively, at a bias voltage (VBIAS) 5524. The control gate voltageof the data control gate voltage 5532 and the reference control gatevoltage 5534 are applied during this time. Once sufficient voltage isdeveloped between the data cell voltage (VCELLD) on the cell bitline5520 and the reference cell voltage on the reference cell bitline 5522,the comparator 5518 is enabled to amplify the difference voltage. As anillustrative example, for a difference current of 0.5 μa, a bitlinecapacitance of 0.5 pF, and a voltage of 10 mV is developed in 10nanoseconds from the relationship that 10 mV equals 0.5 μa times 10nanoseconds divided by 0.5 pF. In this embodiment, no load, such aspullup, is needed for the sensing circuitry. In another embodiment, thedata cell voltage (VCELLD) 5520 and the reference cell voltage (VCELLR)5522 go in a positive direction during the signal development periodinstead of a negative direction as shown in FIG. 56. Voltage modesensing or current mode sensing may be utilized for memory cells 5502and 5504.

FIG. 57 is a schematic diagram illustrating a dynamic charge senseamplifier 5700.

The dynamic charge sense amplifier 5700 comprises a data memory cell5502, a reference memory cell 5504, NMOS transistors 5506, 5508, 5510,5512, 5514 and 5516 arranged in a similar manner as the dynamic senseamplifier 5500 of FIG. 55. The dynamic charge sense amplifier 5700further comprises a plurality of capacitors 5701 through 5704, acomparator 5706, and a plurality of switches 5708 and 5710. Thecapacitor 5701 couples the cell voltage line 5520 to a positive input ofthe differential comparator 5706. The capacitor 5702 couples thereference cell voltage 5522 to a negative input of the comparator 5706.The switches 5708 and 5710 couple respective inverted and non-invertedoutputs 5712 and 5713 to the positive and negative inputs of thecomparator 5706. The capacitors 5703 and 5704 are coupled in parallel tothe switches 5708 and 5710, respectively. An autozero signal 5716 isapplied to the switches 5708 and 5710 and is an active low to enablevoltage signal development of the comparator 5706. In one embodiment,the autozero signal 5716 is buffered from, and thus is logically thesame as, the initialized bitline signal 5526. The negative and positivenodes of the comparator 5706 are initialized or autozeroed at a bias inan autozero state. The capacitors 5703 and 5704 cause the comparator5706 to function as a gain amplifier with a gain equal to the ratio ofthe capacitances of the capacitors 5701 and 5703. In another embodiment,the dynamic charge sense amplifier 5700 does not include the capacitors5703 and 5704, and the comparator 5706 function as a comparator with thecapacitor 5701 and 5702 providing capacitive coupling.

The systems described above may be used for Inverse Voltage ModeSensing, No Current (Digital) Multilevel Mode Sensing, or InverseCurrent Mode Sensing with appropriate modification, and may include anautozero function of the sense amplifier. The autozero function mayinclude equalizing the input and output of the sense amplifier beforesensing or the storage of the reference cell sensing before the datacell sensing to reduce signal path mismatch.

FIG. 58 is a flow diagram illustrating a single bit current sensingbinary search. During the binary search, the data value of a cell beingread is analyzed one bit at a time. As an illustrative example, athree-bit data memory cell for an eight value memory cell is describedin which the bits are B2, B1, and B0 with B2 being the most significantbit and bit B0 being the least significant bit. As an overview of thebinary search, the data cell is set into a current sensing condition andthe sensed data current is compared to a reference current from thereference memory cell. As part of the binary search, the full range ofcurrent values is divided into half and the data cells determine whetherthe data value is in the upper or lower half of the voltage range. Afterthis determination, the selected one-half current range is divided intotwo one-quarter current ranges, and the data current is analyzed todetermine which one-quarter range the data value is in. The one-quarterrange is then divided into half and the data value is analyzed todetermine which of the one-eighth ranges the data value is in, andlikewise for each additional bit. For a three-bit data cell, three suchdeterminations are made. For an n bit data cell, a number ndeterminations are made during the single bit current sensing binarysearch. The eight values can be arbitrary values.

The data cells are set into a sensing condition, and the data memorycell (IDAT) bitline is set into an autozero condition and the data cellsare read (block 5802). The data range being evaluated is divided intohalf and the data current (IDAT) is analyzed to determine whether thedata current is in the upper or lower half of the current range. Thedata current (IDAT) is compared to a reference current from thereference memory cells corresponding to the mid-point of the entire datarange (block 5804). For example, for a three-bit system, the datacurrent (IDAT) is compared to the reference current for the fourthmemory level (IR4). If the sensed data current (IDAT) is greater than orequal to the fourth level reference current (IR4), the first bit beingdetected B2 is set to a high value (B2=1) (block 5806). The data current(IDAT) is in the upper half of the data range, and the upper half of thedata range is divided in half or into two one-quarter data ranges. Thedata current (IDAT) is compared to the midpoint reference current of theupper half which in the illustrative example corresponds to the sixthreference current (IR6). If the data current (IDAT) is greater than orequal to the sixth reference current (IR6) (block 5808), the second databit, B1, is set high (B1=1) (block 5810), and the data current (IDAT) isin the upper half of the data range. The upper quarter of the data rangeis again divided into half and the data current (IDAT) is compared tothe bit level current of the upper quarter range, which is the seventhreference current (IR7). If the data current (IDAT) is greater than orequal to the seventh reference current (IR7) (block 5812), the third bitB0 is set high (B0=1) (block 5814). Thus in this case, the data of thecell corresponds to B2, B1 and B0 equals ‘111’. Otherwise if the datacurrent (IDAT) is less than the seventh reference current (IR7) (block5812), the data current (IDAT) is in the bottom half of the upperquarter of the data range and the last bit B0 is set low (B0=0) (block5816), and the data in the cell corresponds to B2B1B0 equal ‘110’.

On the other hand, if the data current (IDAT) is less than the sixthreference current (IR6) (block 5808), the data is in the quarter rangethat is in the bottom half of the top half of the voltage range and thesecond bit B1 is set low (B1=0) (block 5818). This quarter range is thendivided into two sections corresponding to one-eighth of the overalldata range and the data current (IDAT) is compared to the fifthreference current (IR5). If the data current (IDAT) is greater than orequal to the fifth reference current (block 5820), the data current(IDAT) is in the top half of the quarter range, and the third data bitB0 is set high (B0=1) (block 5822), and the data in the cell correspondsto B2B1B0 equals ‘101’. Otherwise, if the data current (IDAT) is lessthan the fifth reference current (block 5820), the third bit B0 is setlow (B0=0) (block 5816), and the data in the cell corresponds to B2B1B0equals ‘100’.

On the other hand, if the data current (IDAT) is less than the fourthreference current (IR4) (block 5804), the data is in the lower half ofthe current range, and the first bit B2 is set low (B2=0) (block 5824).The half range is divided into two halves corresponding to one-quarterranges of the overall data range, and the data current (IDAT) iscompared to the second reference current (IR2). If the data current(IDAT) is greater than or equal to the second reference current (block5826), the data current (IDAT) is in the upper quarter range of thebottom half range, and the second data bit B1 is set high (B1=1) (block5828). Again, the quarter range is divided into one-eighth ranges andthe data current (IDAT) is compared to the third reference current(IR3). If the data current (IDAT) is greater than or equal to the thirdreference current (block 5830), the data current (IDAT) is in the tophalf of this quarter range, and the third data bit B0 is set high (B0=1)(block 5832), and the data in the cell corresponds to B2B1B0 equals‘011’. Otherwise, if the data current (IDAT) was less than the thirdreference current (block 5830), the third bit B0 is set low (B0=0)(block 5834), and the data in the cell corresponds to B2B1B0 equals‘010’.

On the other hand, if the data current (IDAT) is less than the seconddata current (IR2) (block 5826), the data is in the quarter range thatis in the bottom half of the range, and the second bit B1 is set low(B1=0) (block 5836). This quarter range is then divided into twosections corresponding to one-eighth of the overall data range and thedata current (IDAT) is compared to the first reference current (IR1). Ifthe data current (IDAT) is greater than or equal to the first referencecurrent (block 5838), the data current (IDAT) is in the top half of thisquarter range, and the third bit B0 is set high (B0=1) (block 5840), andthe data of the cell corresponds to B2B1B0 equals ‘001’. Otherwise ifthe data current (IDAT) is less than the first reference current (block5838), the third bit B0 is set low (B0=0) (block 5834), and the data inthe cell corresponds to B2B1B0 equals ‘000’.

FIG. 59 is a flow diagram illustrating a multiple bit current sensingbinary search. The data cell may be connected to multiple senseamplifiers for comparing to different reference currents at the sametime to reduce the number of steps of the search. During a first searchstage, the data current (IDAT) from the data memory cell is applied to aplurality of comparators, which each determine the relationship betweenthe data current and a reference level. One of the comparators comparesthe data current to a reference level that is in the middle of the datarange. The results of this comparison are used to determine whichreference levels are applied to the comparators during a second searchstage. More particularly, the reference levels in the determined datarange are applied. Further, the results of some of the datadetermination are discarded.

As an illustrative example, the data cell is a three-bit data cell thatstores bits B2B1B0 in a manner similar to that described above for FIG.58. Further to the illustrative example, two bits are determined percomparison stage. The three-bit system has eight data values withcorresponding eight reference values. In this illustrative example,three comparators are used to compare the data current to the threedifference reference levels. In the first comparison, the full range isdivided into eight ranges.

The data cells are put into a current sensing condition and the datamemory cell (IDAT) bitline is set into an autozero condition and thedata cells are read (block 5902). The data current (IDAT) is compared toa fourth reference level (IR4) (block 5904), a sixth reference current(IR6) (block 5906), and a second reference current (IR2) (block 5908).The comparison of the data current (IDAT) to the fourth reference level(IR4) determines whether the first data bit B2 is set high or low. Ifthe data current (IDAT) is greater than or equal to the fourth referencecurrent (IR4) (block 5904), the first data bit B2 is set high (B2=1)(block 5910), or otherwise the first data bit B2 is set low (B2=0)(block 5912). The data current (IDAT) is compared to the sixth referencecurrent (IR6) to determine the second data bit B1. If the data current(IDAT) is greater than or equal to the sixth reference current (IR6)(block 5906), the second data bit B1 is set high (B1=1) (block 5914), orotherwise the second data bit B1 is set low (B1=0) (block 5924). Thedata current (IDAT) is compared to the second reference current (IR2).If the data current (IDAT) is greater than or equal to the secondreference current (IR2) (block 5908), the second data bit B1 is set high(B1=1) (block 5928), or otherwise the second data bit B1 is set low(B1=0) (block 5938). If the comparison at block 5904 determines that thedata current (IDAT) is greater than or equal to the fourth referencecurrent, then the data from the comparison at block 5908 is discarded(block 5934). On the other hand, if the data current (IDAT) is less thanthe fourth reference current (IR4) (block 5904), the comparison of block5906 is discarded (block 5922).

During the second stage, the third data bit B0 is determined. If thedata current (IDAT) comparison to the sixth reference current (IR6)(block 5906) indicates the second bit B1 is set high (B1=1) (block5914), the second comparison operation compares the data current (IDAT)to the seventh reference current (IR7). If the data current (IDAT) isgreater than or equal to the seventh reference current (IR7) (block5916), the third data bit B0 is set high (B0=1) (block 5918), and thedata in the cell corresponds to B2B1B0 equals ‘111’. On the other hand,if the data current (IDAT) is less than the seventh reference current(IR7) (block 5916), the third data bit B0 is set low (B0=0) (block5920), and the data in the cell corresponds to B2B1B0 equals ‘110’.

On the other hand, if the data current (IDAT) comparison to the sixthreference current IR6 (block 5906) indicates the second bit B1 is setlow (B1=0) (block 5925), the second comparison operation compares thedata current (IDAT) to the fifth reference current (IR5). If the datacurrent (IDAT) is greater than or equal to the fifth data current (block5924), the third data bit B0 is set high (B0=1) (block 5926), and thedata in the cell corresponds to B2B1B0 equals ‘101’. On the other hand,if the data current (IDAT) is less than the fifth reference current(IR5) (block 5924), the third data bit B0 is set low (B0=0) (block5920), and the data in the cell corresponds to B2B1B0 equals ‘100’.

If the data current comparison to the second reference current IR2indicates the second bit B1 is set high (B1=1) (block 5928), the secondcomparison operation compares the data current to the third referencecurrent IR3. If the data current (IDAT) is greater than or equal to thethird reference current IR3 (block 5930), the third bit B0 is set high(B0=1) (block 5932), and the data in the cell corresponds to B2B1B0equals ‘011’. On the other hand, if the data current (IDAT) is notgreater than or equal to the reference current IR3 (block 5930), thethird bit B0 is set low (B0=0) (block 5936), and the data in the cellcorresponds to B2B1B0 equals ‘010’.

On the other hand, if the data current (IDAT) comparison to the secondreference current IR2 (block 5908) indicates a second bit B1 is set low(B1=0) (block 5938), the second comparison operation compares the datacurrent (IDAT) to the first reference current (IR1). If the data current(IDAT) is greater than or equal to the first data current (IR1) (block5940), the third data bit B0 is set high (B0=1) (block 5942), and thedata in the cell corresponds to B2B1B0 equals ‘001’. On the other hand,if the data current (IDAT) is less than the first reference current(IR1) (block 5940), the third data bit B0 is set low (B0=0) (block5936), and the data in the cell corresponds to B2B1B0 equals ‘000’.

By increasing the number of comparisons done at one time to therebydetermine multiple bits in one comparison cycle, the number ofsequential comparison cycles may be reduced to increase the binarysearch operation.

FIG. 60 is a block diagram illustrating a memory system 6000 includingbuilt-in concurrent byte redundancy.

The memory system 6000 may be a modified version of the super-highdensity, non-volatile multilevel memory integrated circuit system ofFIG. 2A. For clarity, FIG. 60 shows only the portions of the memory thatare different. The memory system 6000 includes a byte decoder 6002, amultiplex column decoder 6004, and a memory array 6006. The byte decoder6002 replaces the byte decoder 152 of FIG. 2A. The memory array 6006 isshown as a single page of the memory array 100. The multiplexer columndecoder 6004 replaces the page select current 120 and the byte selectcircuit 140. As an illustrative example, a page of 512 bytes isdescribed, but other sizes of pages may be used.

The memory array 6006 includes memory cells organized as a normal dataregion 6008, a redundant data region 6010, and a bad byte locator 6012.As an illustrative example, the memory array 6006 comprises 512 bytes ofnormal data in the normal data region 6008, one redundant byte in theredundant data region 6010, and 10 bits for the bad byte locator 6012.The bad byte locator 6012 includes an indicator of whether a bad byteexists in the normal data region 6008 of the memory array 6006, andincludes an address of the location of the bad byte. In the illustrativeembodiment, the bad byte locator 6012 includes one bit for the indicatorand nine bits for the address of the bad byte. The byte decoder 6002decodes and address 6014, which is shown illustratively in FIG. 60 as anaddress 6016 for the 512 bytes of normal data, an address 6018 for oneredundant byte and an address 6020 for 10-bit bad byte locator, andaddresses the page of the memory array 6006, and applies the decodedaddress to the multiplexer column decoder 6004 to address the memoryarray 6006.

In one embodiment, the bad byte latch may be configured to store apredetermined data pattern, such as ‘FF’ to disable programming of thebad byte of the normal data region 6008.

During byte loading, the byte decoder 6002 is addressed by the address6014 to address the page of the memory array 6006 for loading data intothe normal data region 6008, the redundant data region 6010, and the badbyte locator 6012. When the bad byte locator 6012 indicates a bad bytein the normal data region 6008, the byte data corresponding to thisaddress is loaded into the redundant byte 6018. In one embodiment, thebad byte latch may remain with a predetermined data pattern, e.g., ‘FF’,to disable programming. In one embodiment, the bad byte locator 6012 isprogrammed during testing at another location, such as the manufacturer,with the bad byte indicator and the byte address of the location in thenormal data region 6008 that is bad.

During a program and verify operation, the byte decoder 6002 addressesall locations of the normal data region 6008 and the redundant dataregion 6010 for writing the data, for example 513 bytes, into the memorycells of the memory array 6006. The bad byte locator 6012 is used by thebyte decoder 6002 to determine whether the normal data region 6008 has abad byte and the address of the bad byte from which the byte decoder6002 determines the data for storing in the redundant data region 6010.In one embodiment, the bad byte is not programmed if an indicator ‘FF’is used to disable the program and verify for that bad byte.

During a read operation, the byte decoder 6002 addresses all addressesin the page of the memory array 6006 to read the normal data region6008, the redundant data region 6010, and the bad byte locator 6012. Inthe illustrative example, all 513 bytes of data and the 10 bits of thebad byte locator are read. The byte decoder 6002 decodes the bad bytelocator 6012 to determine whether the byte redundancy is invoked and theaddress of the bad byte of the normal data region 6008. If the byteredundancy is invoked, the address of the bad byte is used to switch thedata from the redundant data region 6010, and ignore the data read fromthe bad byte of the normal data region 6008.

During an erase operation, the byte decoder 6014 addresses all memorycells in the normal data region 6008 and the redundant data region 6010.In the illustrative embodiment, the 513 bytes are erased, but the badbyte locator 6012 is not erased. In another embodiment, the eraseincludes a process of storing the data in the bad byte locator 6020 in alatch, erasing the entire memory array 6006 and rewriting the datastored in the latch into the bad byte locator 6020.

In another embodiment, the memory cells may be verified and read indifferent sensing modes. For example, the memory cell may be verified byplacing the memory cell in a voltage mode while reading of the memorycell may be done in a current sensing mode.

In another embodiment, additional reference currents may be formed byinterpolating or extrapolating the values stored in the reference memorycells. For example, the reference memory cells may store data in 0.1 μaincrements in a range from 0.0 to 1.6 μa. A reference current may beinterpolated from currents with adjacent values stored in the referencememory cells, such as by forming a reference current as an average valuebetween adjacent reference values. For example, the first memory cellmay store 0.1 μa and a second memory cell may store 0.2 μa. A referencelevel of 0.15 μa may be generated by dividing the memory range into two.A reference current outside the range may be formed by extrapolation.

In the foregoing description, various methods and apparatus, andspecific embodiments are described. However it should be obvious to theone conversant in the art, various alternatives, modifications, andchanges may be possible without departing from the spirit and the scopeof the invention which is defined by the metes and bounds of theappended claims.

1. A data storage system comprising: a plurality of memory arrays, each memory array comprising: a plurality of memory subarrays, each memory subarray including a plurality of memory cells, selected memory cells being placed in a current sensing mode during a read or verify operation, and a plurality of autozero current sense amplifiers, each autozero sense amplifier being coupled to a group of said memory subarrays, the autozero sense amplifier reading the contents of the memory cells within the corresponding group of memory subarrays.
 2. The data storage system of claim 1 wherein the memory cells are multilevel memory cells.
 3. The data storage system of claim 2 wherein the memory cells are nonvolatile memory cells.
 4. A data storage system comprising: a plurality of memory arrays, each memory array comprising: a plurality of memory subarrays, each memory subarray including a plurality of memory cells, selected memory cells being placed in a current sensing mode during a read or verify operation, and a plurality of autozero local sense amplifiers, each autozero local sense amplifier being disposed adjacent to and coupled to a group of said memory subarrays, the autozero local sense amplifier reading the contents of the memory cells within the corresponding group of memory subarrays.
 5. The data storage system of claim 4 wherein the memory subarrays further comprise a plurality of autozero global sense amplifiers, each autozero global sense amplifier being coupled to a group of said autozero local sense amplifiers.
 6. The data storage system of claim 4 wherein the memory cells are multilevel memory cells.
 7. The data storage system of claim 6 wherein the memory cells are nonvolatile memory cells.
 8. The data storage system of claim 4 wherein the memory cells are nonvolatile memory cells.
 9. A data storage system comprising: a plurality of memory subarrays, each memory subarray comprising a plurality of data memory cells, a plurality of reference memory cells and a plurality of subarray bitlines, each memory cell being configurable to store one of the plurality of signal levels and being coupled to one of said bitlines; a plurality of local sense amplifiers, each of said plurality of local sense amplifiers being coupled to a corresponding one of said plurality of memory subarrays and being selectively coupled to said subarray bitlines in said memory subarray; a plurality of global bitlines, each local sense amplifier being coupled to one of said plurality of global bitlines, the plurality of global bitlines being arranged in a crossed configuration; and a plurality of global sense amplifiers, each global sense amplifier being coupled to a group of said global bitlines.
 10. The data storage system of claim 9 wherein the memory cells are nonvolatile memory cells.
 11. The data storage system of claim 10 wherein the memory cells are multilevel memory cells.
 12. The data storage system of claim 9 wherein the memory cells are multilevel memory cells.
 13. A data storage system comprising: a plurality of memory arrays, each memory array comprising: a plurality of memory subarrays, each memory subarray including a plurality of data memory cells and a plurality of reference memory cells, a plurality of autozero local sense amplifiers, each autozero local sense amplifier being disposed adjacent to and coupled to a group of said memory subarrays, the autozero local sense amplifier reading the contents of the memory cells within the corresponding group of memory subarrays, and a plurality of autozero global sense amplifiers, each autozero global sense amplifier being coupled to a group of said autozero local sense amplifiers.
 14. The data storage system of claim 13 wherein the plurality of memory subarrays are arranged in segments.
 15. The data storage system of claim 13 wherein the memory cells are multilevel memory cells.
 16. The data storage system of claim 13 wherein the memory cells are nonvolatile memory cells.
 17. The data storage system of claim 16 wherein the nonvolatile memory cells are multilevel memory cells.
 18. A data storage system comprising: a plurality of memory arrays, each memory array comprising: a plurality of memory subarrays, each memory subarray including a plurality of memory cells, a plurality of autozero local sense amplifiers, each autozero local sense amplifier being disposed adjacent to and coupled to a group of said memory subarrays, the autozero local sense amplifier reading the contents of the memory cells within the corresponding group of memory subarrays, and a plurality of autozero global sense amplifiers, each autozero global sense amplifier being coupled to a group of said autozero local sense amplifiers, wherein the plurality of memory subarrays are arranged in segments, wherein the segments include data memory cells and reference memory cells.
 19. A data storage system comprising: a plurality of memory arrays, each memory array comprising: a plurality of memory subarrays, each memory subarray including a plurality of memory cells, a plurality of autozero local sense amplifiers, each autozero local sense amplifier being disposed adjacent to and coupled to a group of said memory subarrays, the autozero local sense amplifier reading the contents of the memory cells within the corresponding group of memory subarrays, and a plurality of autozero global sense amplifiers, each autozero global sense amplifier being coupled to a group of said autozero local sense amplifiers, wherein the autozero local sense amplifier is a current sensing autozero local sense amplifier, and the autozero global sense amplifier is a voltage sensing sense amplifier.
 20. A data storage system comprising: a plurality of memory arrays, each memory array comprising: a plurality of memory subarrays, each memory subarray including a plurality of memory cells, a plurality of autozero local sense amplifiers, each autozero local sense amplifier being disposed adjacent to and coupled to a group of said memory subarrays, the autozero local sense amplifier reading the contents of the memory cells within the corresponding group of memory subarrays, and a plurality of autozero global sense amplifiers, each autozero global sense amplifier being coupled to a group of said autozero local sense amplifiers, wherein the autozero local sense amplifier is a current sensing sense amplifier and the global sense amplifier is a current sensing sense amplifier.
 21. A data storage system comprising: a plurality of memory arrays, each memory array comprising: a plurality of memory subarrays, each memory subarray including a plurality of memory cells, a plurality of autozero local sense amplifiers, each autozero local sense amplifier being disposed adjacent to and coupled to a group of said memory subarrays, the autozero local sense amplifier reading the contents of the memory cells within the corresponding group of memory subarrays, and a plurality of autozero global sense amplifiers, each autozero global sense amplifier being coupled to a group of said autozero local sense amplifiers, wherein the local autozero sense amplifier is a voltage sensing sense amplifier, and the global sense amplifier is a current sensing sense amplifier.
 22. A data storage system comprising: a plurality of memory arrays, each memory array comprising: a plurality of memory subarrays, each memory subarray including a plurality of memory cells, a plurality of autozero local sense amplifiers, each autozero local sense amplifier being disposed adjacent to and coupled to a group of said memory subarrays, the autozero local sense amplifier reading the contents of the memory cells within the corresponding group of memory subarrays, and a plurality of autozero global sense amplifiers, each autozero global sense amplifier being coupled to a group of said autozero local sense amplifiers, wherein the plurality of memory subarrays are arranged in pages and each page includes data memory cells and reference memory cells.
 23. A data storage system comprising: a plurality of memory arrays, each memory array comprising: a plurality of memory subarrays, each memory subarray including a plurality of memory cells, a plurality of autozero local sense amplifiers, each autozero local sense amplifier being disposed adjacent to and coupled to a group of said memory subarrays, the autozero local sense amplifier reading the contents of the memory cells within the corresponding group of memory subarrays, and a plurality of autozero global sense amplifiers, each autozero global sense amplifier being coupled to a group of said autozero local sense amplifiers, wherein the plurality of memory subarrays include memory cells arranged in rows, a group of said memory cells being reference memory cells.
 24. A memory system comprising: a plurality of differential sense amplifiers, each differential sense amplifier having first and second inputs, and having an output for generating a data signal indicative of a comparison between signals applied to said first and second inputs; and a plurality of memory segments, each memory segment including a plurality of columns of memory cells, a plurality of column switches, a plurality of reference switches, and a segment switch, each column switch coupled between a corresponding column and the segment switch, the segment switch coupled between the column switches and the first input of the differential sense amplifier, each reference switch being coupled between a corresponding column of memory cells and the second input of a corresponding one of said differential sense amplifiers, one of said memory segments storing reference valves in a group of said columns of memory cells.
 25. The memory system of claim 24 wherein the reference switches of said one of said memory segments storing reference values are enabled and the column switches of another one of said memory segments are enabled to read, program, or verify memory cells in said another one of said memory segments. 